Shift register circuit, display panel, and electronic apparatus

ABSTRACT

Disclosed herein is a shift register circuit that is formed on an insulating substrate with thin film transistors having channels of the same conductivity type and includes shift stages, each of the shift stages including: a first thin film transistor; a second thin film transistor; a 3(1)-th thin film transistor; a 3(2)-th thin film transistor; a 4(1)-th thin film transistor; a 4(2)-th thin film transistor; a fifth thin film transistor; and a sixth thin film transistor.

CROSS REFERENCES TO RELATED APPLICATIONS

This application is a Continuation of application Ser. No. 14/246,810,filed on Apr. 7, 2014, which is a Continuation of application Ser. No.14/077,959, filed on Nov. 12, 2013, now U.S. Pat. No. 8,730,145, issuedon May 20, 2014, which is a Continuation of application Ser. No.13/858,570, filed on Apr. 8, 2013, now U.S. Pat. No. 8,629,828, issuedon Jan. 14, 2014, which is a Divisional application Ser. No. 12/379,526,filed Feb. 24, 2009, now U.S. Pat. No. 8,436,800, issued on May 7, 2013,which claims priority from Japanese Application Number 2008-096716,filed Apr. 3, 2008, the entire contents of which are incorporated hereinby reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention to be described in this specification relates to a shiftregister circuit formed on an insulating substrate with thin filmtransistors including channels of the same conductivity type. Theinvention to be described in this specification has modes as a shiftregister circuit, a display panel, and electronic apparatus.

2. Description of the Related Art

A low-temperature poly-silicon (LIPS) process allows circuit formationwith both an NMOS thin film transistor (TFT) and a PMOS thin filmtransistor. Therefore, it is general to manufacture a circuit (so-calledCMOS circuit) with these two kinds of thin film transistors in thelow-temperature poly-silicon process.

However, the manufacturing of the CMOS circuit inevitably involvesincrease in the number of steps because two kinds of thin filmtransistors are used. This increase in the number of steps causes thelowering of the productivity and increase in the manufacturing cost.

Therefore, it is desired that, if possible, a circuit having the samefunctions as those of the CMOS circuit is achieved with merely thin filmtransistors including channels of the same conductivity type (withmerely NMOS or PMOS thin film transistors) even when the poly-siliconprocess is used.

This kind of same-conductivity-type channel circuit can be applied alsoto circuit formation with amorphous silicon or an organic semiconductor.

For example, with amorphous silicon, circuits other than ones includingmerely NMOS thin film transistors may not be manufactured. As for theorganic TFT, circuits other than ones including merely PMOS thin filmtransistors may not be manufactured.

Because of this background, it is desired to achieve a circuit havingthe same functions as those of the CMOS circuit with merely thin filmtransistors including channels of the same conductivity type (withmerely NMOS or PMOS thin film transistors).

In this specification, attention will be paid on a shift registercircuit particularly. It is obvious that the shift register circuit is ageneral-purpose circuit incorporated in a wide variety of circuits.Therefore, the shift register circuit is not limited to the specificapplication basically. However, the following description is based onthe premise of application to a drive circuit for driving a displaypanel, for convenience.

The following description will deal with a related-art example of ashift register circuit used for an active-matrix driven organic EL(electroluminescence) panel.

FIG. 1 shows a system configuration example of an organic EL panel. Inan organic EL panel 1 shown in FIG. 1, a pixel array part 3, a signalline driver 5, a first control line driver 7, and a second control linedriver 9 are disposed on a panel substrate.

In the pixel array part 3, sub-pixels 11 are arranged in a matrixcorresponding to the display resolution. FIGS. 2 and 3 show examples ofthe equivalent circuit of the sub-pixel 11. In both the circuit examplesof the sub-pixel 11 shown in these diagrams, all of thin filmtransistors are formed of NMOS thin film transistors.

In the diagrams, N1 denotes a sampling transistor, N2 denotes a drivetransistor, N3 denotes a lighting control transistor, and Cs denotes ahold capacitor. Furthermore, WSL, LSL, and PSL denote a write controlline, a lighting control line, and a current supply line, respectively.

FIG. 2 corresponds to a circuit configuration in the case of employing adrive system that achieves the lighting operation and lighting-stopoperation of an organic EL element OLED through the on/off-control ofthe lighting control transistor N3.

On the other hand, FIG. 3 corresponds to a circuit configuration in thecase of employing a drive system that achieves the lighting operationand lighting-stop operation of the organic EL element OLED throughpotential change of the lighting control line LSL. In the configurationof FIG. 3, the lighting control line LSL functions also as the currentsupply line.

FIG. 4 shows a timing chart of writing of a signal potential Vsig (Data)to the sub-pixel 11 shown in FIGS. 2 and 3. FIG. 4A shows the drivewaveform of a signal line DTL. The signal potential Vsig dependent onthe pixel grayscale Data is supplied to the signal line DTL. Themagnitude of a drive current supplied through the drive transistor N2depends on the magnitude of the signal potential Vsig. The organic ELelement OLED is a current-driven element, and larger drive currenttherefor provides higher luminance thereof.

FIG. 4B shows the drive waveform of the write control line WSL. Duringthe period of the H level thereof, the sampling transistor N1 is turnedon and the potential of the signal line DTL is written to the gateelectrode of the drive transistor N2.

FIG. 4C shows the drive waveform of the lighting control line LSL. Thelighting control line LSL is driven with binary values of the H leveland the L level. By this potential switching, the lighting andlighting-stop of the organic EL element OLED are switch-controlled.

The sub-pixel 11 shown in FIG. 2 is different from that shown in FIG. 3in the control amplitude of the lighting control line LSL. This isbecause it is sufficient for the lighting control line LSL to be capableof driving the lighting control transistor N3 in the circuit of FIG. 2,whereas it is necessary for the lighting control line LSL to supply theoperating voltage of the drive transistor N2 and the organic EL elementOLED in the circuit of FIG. 3.

As shown in FIG. 4, after the end of the writing of the signal potentialVsig, the lighting of the organic EL element OLED is achieved when thelighting control line LSL is at the H level, whereas the lighting of theorganic EL element OLED is stopped when the lighting control line LSL isat the L level.

The peak luminance level can be controlled by varying the ratio (Duty)of the lighting period to the one-field period.

The lighting control line LSL (FIG. 4C) is used also for adjustment ofthe moving-image characteristics. For the adjustment of the moving-imagecharacteristics, it is desired to adjust the number of times of lightingin a one-field period and the timing of the lighting period.

Therefore, it is desired for the second control line driver 9 to becapable of outputting plural kinds of pulses.

In addition, in the case of an active-matrix drive system and a generalline-sequential writing system, these pulse waveforms need to beline-sequentially transferred.

That is, it is desired for this kind of control line driver to have thefollowing two functions: the function capable of freely designing thepulse length of the control pulse; and the function capable ofline-sequentially transferring the control pulse to the next stage.

In the sub-pixel 11 shown in FIGS. 2 and 3, for the above-describedoperation of writing the signal potential Vsig, threshold correctionoperation and mobility correction operation for the drive transistor N2are carried out in some cases. FIG. 5 shows a timing chart of thesub-pixel 11 corresponding to FIG. 2. FIG. 6 shows a timing chart of thesub-pixel 11 corresponding to FIG. 3. The difference between thesub-pixel 11 shown in FIG. 2 and that shown in FIG. 3 is whether or notinitialization operation and light-emission period control are separatedfrom each other.

For the light-emission period control, operation of varying the ratio(Duty) of the light-emission period to the lighting-stop period isdesired in order to adjust the peak luminance. Furthermore, for thelight-emission period control, operation of changing the number of timesof switching between the light-emission period and the lighting-stopperiod in the one-field period is desired in order to adjust themoving-image displaying characteristics. For these purposes, the circuitconfiguration of the second control line driver 9 is generally complex.

Therefore, the circuit configuration of FIG. 2, in which the supply linefor the initialization pulse and the supply line for the lighting-periodcontrol pulse, different in the transfer timing of the output pulse, areseparately provided, is advantageous in simplification of the controlinterface. However, as shown in FIG. 2, three control lines are desired:the write control line WSL, the lighting control line LSL, and thecurrent supply line PSL.

In the following, the control operation of the sub-pixel 11, includingthe threshold correction operation, the mobility correction operation,and the light-emission period control, will be described below for thecase of the pixel circuit shown in FIG. 3. Therefore, the descriptionwill be made with reference to FIG. 6.

As described above, the control operation used in the pixel circuitshown in FIG. 2 is the same as that used in the pixel circuit shown inFIG. 3 except for the separation of the initialization operation fromthe light-emission period control.

FIG. 6A shows the drive waveform of the write control line WSL. Forexample, during the period of the H level thereof, the samplingtransistor N1 is turned on and the potential of the signal line DTL iswritten to the gate electrode of the drive transistor N2.

The first H-level period in FIG. 6 is used to correct variation in thethreshold voltage Vth of the drive transistor N2.

On the other hand, the second H-level period in FIG. 6 is used to writethe signal potential Vsig dependent on the pixel grayscale and correctvariation in the mobility p of the drive transistor N2.

The purpose of the inclination of the falling waveform of the pulsecorresponding to the second H-level period is to set the optimummobility correction period for all of the grayscales from thehighest-luminance grayscale (highest signal potential) to thelowest-luminance grayscale (lowest signal potential).

The mobility correction refers to operation for correcting mobilitydifference between the drive transistor N2 having higher mobility p andthe drive transistor N2 having lower mobility p, and the correction timethereof is determined depending on the length of the H-level period ofthe write control line WSL. As this correction period, a longer periodis necessary for lower luminance (lower signal potential) in principle.

FIG. 6B shows the drive waveform of the signal line DTL. Two kinds ofpotentials are applied to the signal line DTL. An offset potential Vofsis used for threshold correction for the drive transistor N2. The signalpotential Vsig is the potential that provides the pixel grayscale. Themagnitude of the drive current supplied through the drive transistor N2depends on the magnitude of the signal potential Vsig. The organic ELelement OLED is a current-driven element, and larger drive currenttherefor provides higher luminance thereof.

FIG. 6C shows the drive waveform of the lighting control line LSL. Thelighting control line LSL is driven with binary values of the H leveland the L level. The first L-level period in FIG. 6 is used to providean initialization period. The second L-level period in FIG. 6 is used toprovide a lighting-stop period after the start of light emission.

The initialization operation is to set the gate-source voltage Vgs ofthe drive transistor N2 higher than the threshold voltage Vth thereof.This operation is necessary before execution of the thresholdcorrection. Hereinafter, this operation will be referred to as thecorrection preparatory operation.

After the correction preparatory operation, the potential of thelighting control line LSL is switched to the H level, with the offsetpotential Vofs continuously applied to the gate electrode of the drivetransistor N2. The operation with this potential relationship is thethreshold correction operation. Upon the start of the thresholdcorrection operation, the source potential Vs of the drive transistor N2gradually increases. The increase in the source potential Vs stops atthe timing when the gate-source voltage Vgs of the drive transistor N2reaches the threshold voltage Vth thereof.

At the end of the writing of the signal potential Vsig, thelight-emission period starts and continues until the next writingperiod. In the light-emission period, the lighting of the organic ELelement OLED is achieved when the lighting control line LSL is at the Hlevel, whereas the lighting of the organic EL element OLED is stoppedwhen the lighting control line LSL is at the L level. The peak luminancelevel can be controlled by varying the ratio of the lighting periodlength in the one-field period.

FIG. 6D shows the potential Vg of the gate electrode of the drivetransistor N2. FIG. 6E shows the potential Vs of the source electrode ofthe drive transistor N2 (the potential of the anode of the organic ELelement OLED).

As described above, the pulse lengths of the write control signal (FIG.6A) and the lighting control signal (FIG. 6C) need to differ dependingon the purpose of the drive operation.

For example, for the former signal, the length of the pulse for thethreshold correction operation needs to be different from that of thepulse for the signal writing and mobility correction operation. For thelatter signal, the length of the pulse corresponding to the period ofthe correction preparatory operation needs to be different from that ofthe pulse for the control of lighting/lighting-stop in thelight-emission period.

Therefore, it is desired for each of the first control line driver 7 andthe second control line driver 9 to be capable of outputting pulses ofplural lengths. In addition, in the case of an active-matrix drivesystem and a general line-sequential writing system, these pulsewaveforms need to be line-sequentially transferred. That is, it isdesired for this kind of control line driver to have the following twofunctions: the function capable of freely designing the pulse length ofthe control pulse; and the function capable of line-sequentiallytransferring the control pulse to the next stage.

FIG. 7 shows a structure example of a shift register circuit that issuitable for use in the control line drive circuit that satisfies theabove-described drive conditions. The shift register circuit shown inFIG. 7 has a configuration obtained by cascade-connecting 2N shiftstages SR(1) to SR(2N). In this shift register circuit, each shift stagecarries out operation of outputting a clock signal ck1 or ck2 inputthereto as an output pulse o(k) to the control line of this shift stageby using, as drive pulses, the output pulses o(k−1) and o(k+1) of othershift stages at the previous and subsequent stages.

FIG. 8 shows drive pulse waveforms of the shift register circuit. Thepulse waveforms of FIG. 8 correspond to the shift register circuitformed with merely NMOS thin film transistors.

FIG. 8A shows a start pulse st for driving the first shift stage, andFIG. 8B shows an end pulse ‘end’ for driving the 2N-th shift stage. FIG.8C shows the clock signal ck1 for the even-numbered shift stages.

FIG. 8D shows the clock signal ck2 for the odd-numbered shift stages.FIG. 8E shows the output pulse o1 of the first shift stage SR(1). FIG.8F shows the output pulse o(k−1) of the (k−1)-th shift stage SR(k−1).Similarly, FIGS. 8G to 8I show the output pulses o of the shift stagescorresponding to the symbols shown in the diagram.

FIG. 9 shows an internal circuit example of the k-th shift stage SR. Inthe circuit of FIG. 9, all of the thin film transistors included in theshift stage SR are NMOS thin film transistors. The output stage of thisshift stage SR is composed of NMOS thin film transistors N11 and N12that are connected in series to each other between a power supply VSSand a clock input terminal ck and NMOS thin film transistors N13 to N16that form a logic gate stage. The connecting midpoint between the thinfilm transistors N11 and N12 is connected to an output node ‘out.’

A complementary capacitor C1 is connected between the gate electrode ofthe thin film transistor N11 and the output terminal ‘out.’ Thiscomplementary capacitor C1 is a capacitor for complementing bootstrapoperation, and is used if the gate capacitance of the thin filmtransistor N11 is insufficient. The complementary capacitor C1 is usedalso as a hold capacitor for a node A equivalent to the controlinterconnect of the thin film transistor N11.

On the other hand, a complementary capacitor C2 is connected between thegate electrode of the thin film transistor N12 and the power supply VSS.The complementary capacitor C2 serves as a hold capacitor for thepotential of a node B equivalent to the control interconnect of the thinfilm transistor N12. The capacitance of the complementary capacitor C2depends on the off-leakage amount of the transistor part and the holdingperiod, and is used if the gate capacitance of the thin film transistorN12 is insufficient.

FIG. 10 shows the potential relationship among the input and outputpulses, the node A, and the node B relating to the shift stage SR shownin FIG. 9. FIG. 10A shows the waveform of the drive pulse input to afirst input terminal in1(k) (the output pulse out(k−1) of the previousregister stage). FIG. 10B shows the waveform of the drive pulse input toa second input terminal in2(k) (the output pulse out(k+1) of thesubsequent register stage).

FIG. 10C shows the waveform of the clock signal ck. FIG. 10D shows thewaveform of the potential of the node A (the gate potential of the thinfilm transistor N11). FIG. 10E shows the waveform of the potential ofthe node B (the gate potential of the thin film transistor N12). FIG.10F shows the waveform of the output pulse that appears at the outputnode ‘out.’

As shown in FIG. 10, the potentials of the node A and the node B areswitched in a complementary manner at each of the timing of the risingof the potential of the first input terminal in1(k) to the H level andthe timing of the rising of the potential of the second input terminalin2(k) to the H level.

This complementary operation is achieved by the function of the logiccircuit composed of the thin film transistors N13 to N16.

For example, when the first input terminal in1(k) is at the H level andthe second input terminal in2(k) is at the L level, the thin filmtransistors N13 and N16 are in the on-state and the thin filmtransistors N14 and N15 are in the off-state. When the first inputterminal in1(k) is at the L level and the second input terminal in2(k)is at the H level, the thin film transistors N14 and N15 are in theon-state and the thin film transistors N13 and N16 are in the off-state.

During the period when the node A is at the H level, the complementarycapacitor C1 and the gate capacitor of the thin film transistor N11 arecharged. Thus, if the clock signal ck is switched to the H level and Vddappears at the output node out(k) in the period during which the node Ais at the H level, the potential of the node A changes in such a manneras to be raised by the potential equal to the voltage charged in thecomplementary capacitor C1 and so on. At this time, a voltage equal toor higher than the threshold voltage Vth(N11) of the thin filmtransistor N11 is ensured as the gate-source voltage Vgs thereof due tobootstrap operation, and thus the waveform having the same amplitude asthat of the clock signal ck appears at the output node out(k).

That is, the shift register circuit shown in FIG. 7 operates to extractthe clock signal ck and output it from the output node sequentially fromthe first register stage. The periods indicated by shaded areas in thewaveform diagrams of FIGS. 10D and 10E correspond to the periods duringwhich the nodes A and B are in the floating state. The floating periodof the node A is indispensable for the bootstrap operation.

Examples of documents about the related art include Japanese PatentLaid-open No. 2005-149624 and Japanese Patent Laid-open No. 2006-277789.

SUMMARY OF THE INVENTION

It can be expected that the shift stage SR having the circuitconfiguration shown in FIG. 9 carries out favorable shift operationbasically. However, in mounting thereof, the influence of off-leakageand coupling needs to be taken into consideration. In particular, thecharacteristics of the thin film transistors vary on aproduction-by-production basis, and the variation in the characteristicsis found even on the same substrate. Therefore, it is desired to achievea shift register circuit that is capable of highly-reliable driveoperation even with this kind of characteristic variation.

Some of the problems that will be involved in the shift stage having thecircuit configuration shown in FIG. 9 will be exemplified below.

Initially, with use of FIGS. 11 and 12, the influence of off-leakagecurrents of the thin film transistors on the holding of the potentialsof the nodes A and B will be described below. FIGS. 11 and 12 show thestate in which the potential of the node B in the floating stategradually decreases toward the L-level potential (VSS) due to theinfluence of off-leakage currents that flow through the thin filmtransistors N13 and N15.

As shown in FIG. 12E, the potential decrease of the node B graduallyprogresses. If the potential of the node B becomes lower than thethreshold voltage Vth(N14) of the thin film transistor N14 in thisprocess, the operation state of the thin film transistor N14 is switchedfrom the on-state to the off-state. The switching of the thin filmtransistor N14 to the off-state precludes the supply of the L-levelpotential (VSS) to the node A, so that the node A is also shifted to thefloating state. In FIG. 12D, the floating state of the node A arisingdue to the leakage currents is indicated by darker shaded areas.

If the node A becomes the floating state in this manner, the followingphenomenon occurs. Specifically, as shown in FIG. 13, the clock signalck that should not be used for this shift stage comes in the node Athrough the diffusion capacitance (coupling capacitance) of thetransistor N11, and fluctuates the potential of the node A toward the Hlevel. This phenomenon will be described below with reference to FIG.14. FIGS. 14A to 14F correspond to FIGS. 12A to 12F, respectively.

In the period during which the node A is in the floating state,pulse-like potential fluctuations occur at the node A as shown by thearrowheads of FIG. 14D. If the amount of this potential change of thenode A is even slightly larger than the threshold voltage Vth(N11) ofthe thin film transistor N11, the thin film transistor N11 is turned on.That is, erroneous operation occurs. As a result, as shown in FIG. 14F,a false pulse waveform appears at the output node ‘out’ in a perioddifferent from the period of the clock signal ck that should be outputoriginally. This false pulse waveform possibly causes erroneoustransferring in the shift register circuit.

The change in the node potential due to off-leakage will occur also atthe node A. FIGS. 15 and 16 show the state in which the potential of thenode A in the floating state is lowered toward the L-level potential VSSdue to the influence of off-leakage currents that flow through the thinfilm transistors N14 and N16.

As shown in FIG. 16D, the occurrence of this kind of off-leakageprecludes the bootstrap operation of the node A. This results ininsufficiency in the gate voltage of the thin film transistor N11, andhence a problem that the output pulse having the maximum amplitude maynot be obtained as shown in FIG. 16F. The waveforms shown in FIGS. 16Ato 16F correspond to those shown in FIGS. 12A to 12F, respectively.

For the shift stage SR having the circuit configuration shown in FIG. 9,anti-load countermeasures specific to the connection structure of theshift register circuit are also desired.

As shown in FIG. 17, the output pulse of the shift stage SR needs todrive not merely the control line corresponding to this shift stage SR(e.g. the write control line WSL, the lighting control line LSL) butalso the first input terminal in1 of the subsequent shift stage SR andthe second input terminal in2 of the previous shift stage SR.

FIG. 18 shows drive pulse waveforms of the shift register circuit. FIG.18A shows a clock signal CKA for the odd-numbered shift stages. FIG. 18Bshows a clock signal CKB for the even-numbered shift stages. FIG. 18Cshows a start pulse ST for driving the first shift stage, and FIG. 18Dshows an output pulse o1 of the first shift stage SR(1). FIG. 18E showsan output pulse o2 of the second shift stage SR(2). FIG. 18F shows anoutput pulse oE of the final shift stage SR(E). FIG. 18G shows an endpulse END for driving the final shift stage.

As above, each shift stage SR is desired to drive both the node B of theprevious shift stage and the node A of the subsequent shift stage, inaddition to the control line corresponding to this shift stage SR.

This characteristic is shown in FIG. 19. FIG. 19 shows the relationshipamong the internal potentials of the shift stages from the first shiftstage to the third shift stage.

FIG. 19A shows the clock signal CKA for the odd-numbered shift stages.FIG. 19B shows the clock signal CKB for the even-numbered shift stages.

FIGS. 19C, 19D, and 19E show the waveforms of the potential of the nodeA, the potential of the node B, and the output pulse, respectively, ofthe first shift stage. FIGS. 19F, 19G, and 19H show the waveforms of thepotential of the node A, the potential of the node B, and the outputpulse, respectively, of the second shift stage.

FIGS. 19I, 19J, and 19K show the waveforms of the potential of the nodeA, the potential of the node B, and the output pulse, respectively, ofthe third shift stage.

As shown by the arrowheads in FIG. 19, the output pulse of the secondshift stage (FIG. 19H) is used to charge the node B of the first shiftstage and the node A of the third shift stage to the H level.

Specifically, the output pulse is used to charge the interconnectcapacitors of the nodes A and B. Therefore, the clock signal CK isdesired to have such drive ability as to drive this high loadcapacitance. Specifically, it is desired to increase the size of thebuffer circuit that supplies the clock signal.

However, the enhancement in the drive ability causes increase in thepower consumption and narrowing of the drive margin at high frequencies.

As described above, the technical problems that should be solved stillremain in the shift stage that has been proposed as the related art.

To address the problems, the present inventors propose a shift registercircuit having a shift stage that allows elimination of at least one ofthese technical problems.

(A) Shift Register Circuit (A-1) FIRST SOLUTION

The present inventors propose to employ the following structures foreach of shift stages that are cascade-connected in a shift registercircuit formed on an insulating substrate with thin film transistorshaving channels of the same conductivity type.

(a) a first thin film transistor configured to have one main electrodeconnected to a clock input terminal and the other main electrodeconnected to an output terminal;(b) a second thin film transistor configured to have one main electrodeconnected to the output terminal and the other main electrode connectedto a first power supply;(c) a 3(1)-th thin film transistor configured to have one main electrodeconnected to the first power supply and the other main electrodeconnected to the control interconnect of the second thin filmtransistor;(d) a 3(2)-th thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the second thin film transistor, and acontrol electrode connected to the control interconnect of the firstthin film transistor;(e) a 4(1)-th thin film transistor configured to have one main electrodeconnected to the first power supply and the other main electrodeconnected to the control interconnect of the first thin film transistor;(f) a 4(2)-th thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the first thin film transistor, and acontrol electrode connected to the control interconnect of the secondthin film transistor;(g) a fifth thin film transistor configured to have one main electrodeconnected to a second power supply, the other main electrode connectedto the control interconnect of the second thin film transistor, and acontrol electrode connected to the control interconnect of the 4(1)-ththin film transistor and a second input terminal; and(h) a sixth thin film transistor configured to have one main electrodeconnected to the second power supply, the other main electrode connectedto the control interconnect of the first thin film transistor, and acontrol electrode connected to the control interconnect of the 3(1)-ththin film transistor and a first input terminal.

In the shift stage according to this solution, drive signals are coupledto merely the control terminal of the 3(1)-th thin film transistor, thecontrol terminal of the 4(1)-th thin film transistor, the controlterminal of the fifth thin film transistor, and the control terminal ofthe sixth thin film transistor. That is, it is sufficient for the drivesignals to have an ability to drive the load capacitors of the controlterminals. Because the capacitance of the load capacitors that should bedriven is low, a shift register circuit that allows high-frequencydriving and reduction in the power consumption simultaneously can beachieved.

Furthermore, in the shift stage according to this solution, the 3(1)-ththin film transistor and the 4(1)-th thin film transistor act to promoteswitching of the control potentials of the first thin film transistorand the second thin film transistor.

In addition, in the shift stage according to this solution, the 3(2)-ththin film transistor and the 4(2)-th thin film transistor act to holdthe control potentials of the first thin film transistor and the secondthin film transistor.

These operations are effective for prevention of leakage in a bootstrapperiod and reduction in through-current. The reduction inthrough-current is effective also for achievement of high-frequencydriving and lower power consumption.

(A-2) SECOND SOLUTION

The present inventors propose to set the signal amplitude of a startpulse and an end pulse smaller than that of a clock signal output fromthe output terminal of each of the shift stages, if the followingconnection forms (a) to (c) are employed for the shift register circuitaccording to the first solution.

(a) for each of the shift stages included in the shift register circuitexcept for the first shift stage and the final shift stage, the outputterminal of the previous shift stage is connected to the first inputterminal and the output terminal of the subsequent shift stage isconnected to the second input terminal.(b) the start pulse is input to the first input terminal of the firstshift stage.(c) the end pulse is input to the second input terminal of the finalshift stage.

In this manner, this solution achieves a decrease in the amplitude ofthe start pulse and the end pulse by utilizing the bootstrap operationof the shift stage. The decrease in the amplitude of the drive pulsesallows reduction in the power consumption.

(A-3) THIRD SOLUTION

The present inventors propose a structure in which control electrodesare formed on both the sides of a channel layer in the first thin filmtransistor in the shift register circuit according to the secondsolution.

In this structure, the gate capacitance of the first thin filmtransistor is increased, which can enhance the bootstrap gain. Thisallows a corresponding further decrease in the amplitude of the startpulse and the end pulse.

(A-4) FOURTH SOLUTION

Furthermore, the present inventors propose a structure in which W/L (Wdenotes the channel width and L denotes the channel length) of the fifththin film transistor is equal to or larger than W/L of the 3(1)-th and3(2)-th thin film transistors in the shift register circuit according tothe above-described solution.

In this structure, the off-leakage amount of the fifth thin filmtransistor is relatively larger than that of the 3(1)-th and 3(2)-ththin film transistors. The fifth thin film transistor is connected tothe second power supply, which supplies the same potential as thepotential that should be held. Thus, potential change of the controlinterconnect, which causes erroneous operation, can be minimized evenwhen characteristic variation remains.

(A-5) FIFTH SOLUTION

In another structure proposed by the present inventors, in the shiftregister circuit according to the above-described solution, the controlelectrode is formed on merely one side of a channel layer in the fifththin film transistor. On the other hand, control electrodes are formedon both the sides of a channel layer in each of the 3(1)-th and 3(2)-ththin film transistors.

In this structure, the off-leakage amount of the fifth thin filmtransistor is relatively larger than that of the 3(1)-th and 3(2)-ththin film transistors. The fifth thin film transistor is connected tothe second power supply, which supplies the same potential as thepotential that should be held. Thus, potential change of the controlinterconnect, which causes erroneous operation, can be minimized evenwhen characteristic variation remains.

(A-6) SIXTH SOLUTION

In another structure proposed by the present inventors, in the shiftregister circuit according to the above-described solution, thesource-shield length of the fifth thin film transistor is smaller thanthat of the 3(1)-th and 3(2)-th thin film transistors.

In this structure, the off-leakage amount of the fifth thin filmtransistor is relatively larger than that of the 3(1)-th and 3(2)-ththin film transistors. The fifth thin film transistor is connected tothe second power supply, which supplies the same potential as thepotential that should be held. Thus, potential change of the controlinterconnect, which causes erroneous operation, can be minimized evenwhen characteristic variation remains.

(A-7) SEVENTH SOLUTION

In another structure proposed by the present inventors, in the shiftregister circuit according to the above-described solution, the lengthof a lightly doped drain (LDD) region of the fifth thin film transistoris smaller than the length of an LDD region of the 3(1)-th and 3(2)-ththin film transistors.

In this structure, the off-leakage amount of the fifth thin filmtransistor is relatively larger than that of the 3(1)-th and 3(2)-ththin film transistors. The fifth thin film transistor is connected tothe second power supply, which supplies the same potential as thepotential that should be held. Thus, potential change of the controlinterconnect, which causes erroneous operation, can be minimized evenwhen characteristic variation remains.

(A-8) EIGHTH SOLUTION

Furthermore, the present inventors propose to employ the followingconnection forms (a) and (b) for the shift register circuit according tothe first solution.

(a) a start pulse is input to the first input terminal of the firstshift stage of the shift stages, and the first input terminal of each ofthe shift stages except the first shift stage is connected to the outputterminal of the previous shift stage.(b) two kinds of clock signals having phases adjacent to each other, ofthree kinds of clock signals having phases different from each other,are supplied to the clock input terminal and the second input terminalof each of the shift stages in such a way that the phase combination ofthese two kinds of clock signals is cyclically shifted on astage-by-stage basis.

The shift register circuit according to this solution operates based onthree kinds of clocks. In this case, the interconnects among the shiftstages can be simplified. Furthermore, because the drive pulse isperiodically input to the second input terminal, the period during whichthe control potential of the second thin film transistor is in thefloating state can be shortened. This is effective for enhancement inthe operation reliability.

(A-9) NINTH SOLUTION

Furthermore, the present inventors propose to employ the followingstructures for each of shift stages that are cascade-connected in ashift register circuit formed on an insulating substrate with thin filmtransistors having channels of the same conductivity type.

(a) a first thin film transistor configured to have one main electrodeconnected to a clock input terminal and the other main electrodeconnected to an output terminal;(b) a second thin film transistor configured to have one main electrodeconnected to the output terminal and the other main electrode connectedto a first power supply;(c) a 3(1)-th thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the second thin film transistor, and acontrol electrode connected to a first input terminal;(d) a 3(2)-th thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the second thin film transistor, and acontrol electrode connected to the control interconnect of the firstthin film transistor;(e) a 4(1)-th thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the first thin film transistor, and acontrol electrode connected to a second input terminal;(f) a 4(2)-th thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the first thin film transistor, and acontrol electrode connected to the control interconnect of the secondthin film transistor;(g) a fifth thin film transistor configured to have one main electrodeconnected to the second power supply and the other main electrodeconnected to the control interconnect of the second thin filmtransistor;(h) a sixth thin film transistor configured to have one main electrodeconnected to the second power supply and the other main electrodeconnected to the control interconnect of the first thin film transistor;(i) a seventh thin film transistor configured to have one main electrodeconnected to the control interconnect of the fifth thin film transistor,the other main electrode connected to the second input terminal, and acontrol electrode connected to the second power supply; and(j) an eighth thin film transistor configured to have one main electrodeconnected to the control interconnect of the sixth thin film transistor,the other main electrode connected to the first input terminal, and acontrol electrode connected to the second power supply.

In the shift stage according to this solution, the control electrodes ofthe first and second thin film transistors can be rapidly charged due tothe bootstrap operation of the control electrodes of the fifth and sixththin film transistors. This makes it possible to enhance the drivefrequency.

Furthermore, the start potentials in the bootstrap operation of thefirst and second thin film transistors can be set higher than those inthe first solution in the case of an NMOS-type shift stage, and can beset lower than those in the first solution in the case of a PMOS-typeshift stage. Thus, the necessary bootstrap amount can be decreased. Thisallows reduction in the capacitance of the capacitor for complementingbootstrap operation and is effective for decreasing the layout area.

(A-10) TENTH SOLUTION

Furthermore, the present inventors propose to employ the followingstructures for each of shift stages that are cascade-connected in ashift register circuit formed on an insulating substrate with thin filmtransistors having channels of the same conductivity type.

(a) a first thin film transistor configured to have one main electrodeconnected to a clock input terminal and the other main electrodeconnected to an output terminal;(b) a second thin film transistor configured to have one main electrodeconnected to the output terminal and the other main electrode connectedto a first power supply;(c) a third thin film transistor configured to have one main electrodeconnected to the first power supply and the other main electrodeconnected to the control interconnect of the second thin filmtransistor;(d) a fourth thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the first thin film transistor, and acontrol electrode connected to the control interconnect of the secondthin film transistor;(e) a fifth thin film transistor configured to have one main electrodeconnected to the second power supply, the other main electrode connectedto the control interconnect of the second thin film transistor, and acontrol electrode connected to a second input terminal; and(f) a sixth thin film transistor configured to have one main electrodeconnected to the second power supply, the other main electrode connectedto the control interconnect of the first thin film transistor, and acontrol electrode connected to the control interconnect of the thirdthin film transistor and a first input terminal.

In the shift stage according to this solution, drive signals are coupledto merely the control terminal of the third thin film transistor, thecontrol terminal of the fifth thin film transistor, and the controlterminal of the sixth thin film transistor. That is, it is sufficientfor the drive signals to have an ability to drive the load capacitors ofthe control terminals. Because the capacitance of the load capacitorsthat should be driven is low, a shift register circuit that allowshigh-frequency driving and reduction in the power consumptionsimultaneously can be achieved.

(A-11) ELEVENTH SOLUTION

Furthermore, the present inventors propose to employ the followingstructures for each of shift stages that are cascade-connected in ashift register circuit formed on an insulating substrate with thin filmtransistors having channels of the same conductivity type.

(a) a first thin film transistor configured to have one main electrodeconnected to a clock input terminal and the other main electrodeconnected to an output terminal;(b) a second thin film transistor configured to have one main electrodeconnected to the output terminal and the other main electrode connectedto a first power supply;(c) a third thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the second thin film transistor, and acontrol electrode connected to the control interconnect of the firstthin film transistor;(d) a fourth thin film transistor configured to have one main electrodeconnected to the first power supply, the other main electrode connectedto the control interconnect of the first thin film transistor, and acontrol electrode connected to the control interconnect of the secondthin film transistor;(e) a fifth thin film transistor configured to have one main electrodeconnected to the second power supply, the other main electrode connectedto the control interconnect of the second thin film transistor, and acontrol electrode connected to a second input terminal; and(f) a sixth thin film transistor configured to have one main electrodeconnected to the second power supply, the other main electrode connectedto the control interconnect of the first thin film transistor, and acontrol electrode connected to a first input terminal.

In the shift stage according to this solution, drive signals are coupledto merely the control terminal of the fifth thin film transistor and thecontrol terminal of the sixth thin film transistor. That is, it issufficient for the drive signals to have an ability to drive the loadcapacitors of the control terminals. Because the capacitance of the loadcapacitors that should be driven is low, a shift register circuit thatallows high-frequency driving and reduction in the power consumptionsimultaneously can be achieved.

In addition, in the shift stage according to this solution, the thirdthin film transistor and the fourth thin film transistor act to hold thepotentials of the control interconnects of the first thin filmtransistor and the second thin film transistor. This operation iseffective for prevention of leakage in a bootstrap period and canenhance the stability of the operation.

(A-12) TWELFTH SOLUTION

Moreover, the present inventors propose that the respective shiftregister circuits according to the above-described solutions areincorporated in a display panel as at least a part of a drive circuitincluded in the display panel.

In addition, the present inventors propose electronic apparatusincluding a display panel having this kind of drive circuit. Theelectronic apparatus includes the display panel, a system controllerthat controls the operation of the entire system, and an operation inputunit that accepts an operation input to the system controller.

The above-described shift register circuits based on thin filmtransistors with channels of the same conductivity type are lesssusceptible to the influence of characteristic variation due tomanufacturing variation and allow higher-speed operation and lower powerconsumption compared with the related-art circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram showing a system configuration example of an organicEL panel;

FIG. 2 is a diagram showing an equivalent circuit of a sub-pixel (NMOStype);

FIG. 3 is a diagram showing an equivalent circuit of the sub-pixel (NMOStype);

FIGS. 4A, 4B, and 4C are diagrams for explaining drive timings of thesub-pixel;

FIGS. 5A, 5B, 5C, 5D, 5E, and 5F are diagrams showing drive waveformscorresponding to FIG. 2;

FIGS. 6A, 6B, 6C, 6D, and 6E are diagrams showing drive waveformscorresponding to FIG. 3;

FIG. 7 is a diagram showing a circuit example of a shift registercircuit (scanner);

FIGS. 8A, 8B, 8C, 8D, 8E, 8F, 8G, 8H, and 8I are diagrams showing drivewaveforms of the shift register circuit (scanner) (NMOS type);

FIG. 9 is a diagram for explaining the internal structure of a shiftstage with a bootstrap function;

FIGS. 10A, 10B, 10C, 10D, 10E, and 10F are diagrams for explaining drivewaveforms in the shift stage;

FIG. 11 is a diagram for explaining off-leakage paths;

FIGS. 12A, 12B, 12C, 12D, 12E, and 12F are diagrams for explaining theinfluence of the off-leakage on the potential held by a node B;

FIG. 13 is a diagram for explaining off-leakage paths and a couplingpath;

FIGS. 14A, 14B, 14C, 14D, 14E, and 14F are diagrams for explaining theinfluence of the coupling on the potential held by a node;

FIG. 15 is a diagram for explaining other off-leakage paths;

FIGS. 16A, 16B, 16C, 16D, 16E, and 16F are diagrams for explaining theinfluence of the off-leakage on the bootstrap operation of a node A;

FIG. 17 is a diagram showing a circuit example of the shift registercircuit (scanner);

FIGS. 18A, 18B, 18C, 18D, 18E, 18F, and 18G are diagrams showing drivewaveforms of the shift register circuit (scanner) (NMOS type);

FIGS. 19A, 19B, 19C, 19D, 19E, 19F, 19G, 19H, 19I, 19J, and 19K arediagrams for explaining the drive load of a shift stage;

FIG. 20 is a diagram showing a system configuration example of anorganic EL panel according to a form example;

FIG. 21 is a diagram showing one example of the connection form of ashift register circuit according to the form example;

FIG. 22 is a diagram showing a form example of a shift stage (NMOStype);

FIGS. 23A, 23B, 23C, 23D, 23E, and 23F are diagrams showing drivewaveforms of the shift stage shown in FIG. 22;

FIG. 24 is a diagram showing an equivalent circuit of a sub-pixel (PMOStype);

FIG. 25 is a diagram showing an equivalent circuit of the sub-pixel(PMOS type);

FIGS. 26A, 26B, and 26C are diagrams showing drive waveforms of theshift register circuit (scanner) (PMOS type);

FIG. 27 is a diagram showing a form example of a shift stage (PMOStype);

FIGS. 28A, 28B, 28C, 28D, 28E, and 28F are diagrams showing drivewaveforms of the shift stage shown in FIG. 27;

FIG. 29 is a diagram for explaining the influence of off-leakage;

FIG. 30 is a diagram showing a form example of a shift stage (NMOStype);

FIGS. 31A, 31B, 31C, 31D, 31E, and 31F are diagrams showing drivewaveforms of the shift stage shown in FIG. 30;

FIG. 32 is a diagram showing a form example of a shift stage (PMOStype);

FIG. 33 is a diagram showing a form example of a shift stage (NMOStype);

FIGS. 34A, 34B, 34C, 34D, 34E, and 34F are diagrams showing drivewaveforms of the shift stage shown in FIG. 33;

FIG. 35 is a diagram showing a form example of a shift stage (PMOStype);

FIG. 36 is a diagram for explaining the influence of off-leakage;

FIG. 37 is a diagram showing a form example of a shift stage (NMOStype);

FIG. 38 is a diagram showing a form example of a shift stage (PMOStype);

FIGS. 39A, 39B, and 39C are diagrams for explaining the gate structuresof thin film transistors;

FIGS. 40A and 40B are diagrams for explaining the relationship betweenthe gate structure and off-leakage;

FIG. 41 is a diagram showing a form example of a shift stage (NMOStype);

FIG. 42 is a diagram showing a form example of a shift stage (PMOStype);

FIGS. 43A and 43B are diagrams for explaining the relationship betweenthe source-shield length and off-leakage (NMOS type);

FIGS. 44A and 44B are diagrams for explaining the relationship betweenthe source-shield length and off-leakage (PMOS type);

FIG. 45 is a diagram showing a form example of a shift stage (NMOStype);

FIG. 46 is a diagram showing a form example of a shift stage (PMOStype);

FIG. 47 is a diagram showing a form example of a shift stage (NMOStype);

FIG. 48 is a diagram showing a form example of a shift stage (NMOStype);

FIGS. 49A, 49B, 49C, 49D, 49E, and 49F are diagrams showing drivewaveforms of the shift stage shown in FIG. 48;

FIG. 50 is a diagram showing the potential relationship necessary forbootstrap operation;

FIGS. 51A, 51B, 51C, 51D, 51E, 51F, and 51G are diagrams showing drivewaveforms of a shift register circuit employing the shift stage shown inFIG. 48;

FIG. 52 is a diagram showing a form example of a shift stage (PMOStype);

FIGS. 53A, 53B, and 53C are diagrams showing the relationship betweenthe gate structure and the gate capacitance of a thin film transistor;

FIG. 54 is a diagram showing a form example of a shift stage (NMOStype);

FIGS. 55A and 55B are diagrams for explaining the relationship betweenthe gate structure and current of a thin film transistor;

FIG. 56 is a diagram showing a form example of a shift stage (PMOStype);

FIG. 57 is a diagram showing one example of the connection form of ashift register circuit according to a form example;

FIGS. 58A, 58B, 58C, 58D, 58E, 58F, and 58G are diagrams showing drivewaveforms of the shift register circuit shown in FIG. 57 (NMOS type);

FIGS. 59A, 59B, 59C, 59D, 59E, and 59F are diagrams showing drivewaveforms of a shift stage when the connection form shown in FIG. 57 isemployed (NMOS type);

FIGS. 60A, 60B, 60C, 60D, 60E, 60F, and 60G are diagrams showing drivewaveforms of the shift register circuit shown in FIG. 57 (PMOS type);

FIGS. 61A, 61B, 61C, 61D, 61E, and 61F are diagrams showing drivewaveforms of a shift stage when the connection form shown in FIG. 57 isemployed (PMOS type);

FIG. 62 is a diagram showing a circuit example of the shift stageaccording to the form example (NMOS type);

FIGS. 63A, 63B, 63C, 63D, 63E, and 63F are diagrams showing drivewaveforms of the shift stage shown in FIG. 62;

FIG. 64 is a diagram showing an equivalent circuit when the thin filmtransistor connected to first and second input terminals is turned on;

FIG. 65 is a diagram for explaining change in a node potential indriving with the equivalent circuit shown in FIG. 64;

FIG. 66 is a diagram for explaining an operation principle;

FIG. 67 is a diagram showing a circuit example of a shift stageaccording to a form example (NMOS type);

FIGS. 68A, 68B, 68C, 68D, 68E, 68F, 68G, and 68H are diagrams showingdrive waveforms of the shift stage shown in FIG. 67;

FIGS. 69A and 69B are diagrams showing an equivalent circuit when thethin film transistor connected to first and second input terminals inthe shift stage of FIG. 67 is turned on;

FIG. 70 is a diagram for explaining change in a node potential indriving with the equivalent circuit shown in FIG. 69;

FIG. 71 is a diagram showing a circuit example of a shift stageaccording to a form example (PMOS type);

FIGS. 72A, 72B, 72C, 72D, 72E, 72F, 72G, and 72H are diagrams showingdrive waveforms of the shift stage shown in FIG. 71;

FIG. 73 is a diagram showing an appearance configuration example of adisplay panel;

FIG. 74 is a diagram showing a functional configuration example ofelectronic apparatus;

FIG. 75 is a diagram showing a commercial product example of theelectronic apparatus;

FIGS. 76A and 76B are diagrams showing a commercial product example ofthe electronic apparatus;

FIG. 77 is a diagram showing a commercial product example of theelectronic apparatus;

FIGS. 78A and 78B are diagrams showing a commercial product example ofthe electronic apparatus; and

FIG. 79 is a diagram showing a commercial product example of theelectronic apparatus.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The following description will deal with an example in which anembodiment of the present invention proposed by this specification isapplied to a drive circuit in an active-matrix driven display panel.

Well-known or publicly-known techniques in the related-art technicalfield are applied to part that is not particularly illustrated ordescribed in the present specification.

It should be noted that the form examples to be described below ismerely one embodiment example of the present invention and the presentinvention is not limited thereto.

(A) System Configuration of Display Panel

The following form example relates to an organic EL panel. FIG. 20 showsa system configuration example of an organic EL panel according to theform example. The same parts in FIG. 20 as those in FIG. 1 are given thesame numerals and symbols.

An organic EL panel 21 according to the form example includes a pixelarray part 3, a signal line driver 5, a first control line driver 23,and a second control line driver 25 on a panel substrate.

A shift register circuit that transfers a drive pulse in the verticaldirection of the display panel is used for the first and second controlline drivers 23 and 25 according to the form example. FIG. 21 shows thebasic circuit configuration of the shift register circuit used for thecontrol line drivers.

In this shift register circuit, each shift stage operates to extract aclock signal ck1 or ck2 input thereto at a predetermined timing by useof the output pulses of the previous and subsequent shift stages asdrive pulses. In the case of the shift register circuit shown in FIG.21, input of a start pulse st to the first shift stage and input of anend pulse ‘end’ to the final shift stage are necessary.

(B) Configuration Example of Shift Register Circuit

The following description will deal with two cases: the case in whichthe shift register circuit proposed by the present inventors is formedwith merely NMOS thin film transistors; and the case in which it isformed with merely PMOS thin film transistors. The connectionrelationship among the respective thin film transistors is the samebetween these two cases, irrespective of the difference in theconductivity type of the semiconductor serving as the channel. However,the potential relationship of the drive waveforms and the input/outputwaveforms is reversed between these two cases.

(B-1) FIRST FORM EXAMPLE OF SHIFT STAGE (a) NMOS Type

FIG. 22 shows a first form example of the shift stage included in theshift register circuit. FIG. 23 shows the corresponding drive waveforms.

The shift stage shown in FIG. 22 is composed of an output stageincluding thin film transistors N21 and N22 and a logic gate stageincluding thin film transistors N23 to N26. This form example is definedby the logic gate stage, which drives the output stage.

Initially, the configuration of the output stage will be describedbelow. The output stage is composed of the thin film transistors N21 andN22 that are connected in series to each other between a first powersupply (lower potential VSS) and a clock input terminal CK. Theconnecting midpoint between the thin film transistors N21 and N22 servesas an output node OUT.

To the clock input terminal CK, the clock signal ck1 or ck2 of FIG. 21is supplied depending on the position of the shift stage. The potentialof the clock signal is either the lower potential VSS or a higherpotential Vdd.

In this form example, a complementary capacitor C1 is connected betweenthe gate electrode of the thin film transistor N21 and the output nodeOUT. This complementary capacitor C1 is a capacitor for complementingbootstrap operation, and is used if the gate capacitance of the thinfilm transistor N21 is insufficient. The complementary capacitor C1 isused also as a hold capacitor for a node A.

On the other hand, a complementary capacitor C2 is connected between thegate electrode of the thin film transistor N22 and the first powersupply (VSS). The complementary capacitor C2 is a hold capacitor for thepotential of a node B. The capacitance of the complementary capacitor C2depends on the off-leakage amount of the thin film transistor N22 andthe potential holding period, and is used if the gate capacitance of thethin film transistor N22 is insufficient.

Next, the configuration of the logic gate stage will be described below.The logic gate stage is composed of the thin film transistors N23 toN26.

For the thin film transistor N23, one main electrode thereof isconnected to the first power supply (lower potential VSS), and the othermain electrode thereof is connected to the control interconnect of thethin film transistor N22 (i.e. to the node B).

The gate electrode of the thin film transistor N23 is connected also toa first input terminal in1. This connection allows the thin filmtransistor N23 to carry out operation of supplying the lower potentialVSS to the node B during the period when the first input terminal in1 isat the H level.

For the thin film transistor N24, one main electrode thereof isconnected to the first power supply (lower potential VSS), and the othermain electrode thereof is connected to the control interconnect of thethin film transistor N21 (i.e. to the node A). The gate electrode of thethin film transistor N24 is connected to the control interconnect of thethin film transistor N22 (i.e. to the node B). This connection allowsthe thin film transistor N24 to carry out operation of supplying thelower potential VSS to the node A during the period when the node B isat the H level.

For the thin film transistor N25, one main electrode thereof isconnected to a second power supply (higher potential Vdd), and the othermain electrode thereof is connected to the control interconnect of thethin film transistor N22 (i.e. to the node B). The gate electrode of thethin film transistor N25 is connected to a second input terminal in2.This connection allows the thin film transistor N25 to carry outoperation of supplying the higher potential Vdd to the node B during theperiod when the second input terminal in2 is at the H level.

For the thin film transistor N26, one main electrode thereof isconnected to the second power supply (higher potential Vdd), and theother main electrode thereof is connected to the control interconnect ofthe thin film transistor N21 (i.e. to the node A). The gate electrode ofthe thin film transistor N26 is connected to the gate electrode of thethin film transistor N23 and the first input terminal in1. Thisconnection allows the thin film transistor N26 to carry out operation ofsupplying the higher potential Vdd to the node A during the period whenthe first input terminal in1 is at the H level.

FIG. 23 shows the drive waveforms of the shift stage according to thisform example. FIG. 23A shows the input waveform of the first inputterminal in1. FIG. 23B shows the input waveform of the second inputterminal in2. FIG. 23C shows the input waveform of the clock signal ck.FIG. 23D shows the drive waveform of the node A. FIG. 23E shows thedrive waveform of the node B. FIG. 23F shows the signal waveform of theoutput pulse that appears at the output node OUT.

As shown in FIG. 23, at the timing when the H-level potential is inputto the first input terminal in1, the potential of the node A rises up tothe H level (=the higher potential Vdd−Vth(N26)) and the potential ofthe node B falls down to the L level (=the lower potential VSS). Thisturns the thin film transistor N21 to the state of being capable ofcapturing the clock signal. During the period when the first inputterminal in1 is at the H level, the gate capacitor of the thin filmtransistor N21 and the complementary capacitor C1 are charged.Therefore, also after the potential of the first input terminal in1 hasfallen down to the L level, the node A and the node B keep thepotentials held immediately before this potential falling of the firstinput terminal in1, in the floating state.

If the clock signal is input in this state, the potential of the node Ais raised by bootstrap operation as shown in FIG. 23D. As a result, thehigher potential Vdd, which is the same as the potential of the clocksignal, appears at the output node OUT. The floating state of the nodesA and B is continued until the H-level potential is input to the secondinput terminal in2.

Upon the input of the H-level potential to the second input terminalin2, the potential of the node B rises up to the H level (the higherpotential Vdd−Vth(N25)). In linkage with this potential change of thenode B, the thin film transistor N24 is switched to the on-state, sothat the potential of the node A is forcibly lowered to the L level(lower potential VSS).

The above-described operation is carried out in each shift stage. Inthis form example, the first and second input terminals in1 and in2should just drive the gate electrodes of the thin film transistors N25and N26. That is, they do not need to drive the nodes A and B(interconnect capacitors) unlike the related-art example shown in FIG.9.

Therefore, the load of the input terminals in1 and in2 is reducedcompared with the related-art example. As a result, the load of thetransfer pulse is reduced. This allows driving at a higher frequencycompared with the shift register circuit according to the related-artexample.

(b) PMOS Type

The above-described NMOS-type shift stage is used if the sub-pixels 11in the pixel array part 3 have the NMOS-type circuit configuration shownin FIG. 2 or FIG. 3.

Therefore, if the sub-pixels 11 in the pixel array part 3 have thePMOS-type circuit configuration shown in FIG. 24 or FIG. 25, a shiftregister circuit including PMOS-type shift stages should be used.

The configurations of the sub-pixel 11 shown in FIG. 24 and FIG. 25 arebasically the same as those of FIG. 2 and FIG. 3, respectively, exceptfor replacement of the respective NMOS thin film transistors in FIG. 2and FIG. 3 by PMOS thin film transistors. Therefore, as shown in FIG.26, the drive waveforms in these configurations are obtained byinterchanging the H level and the L level of the write control line WSLand the lighting control line LSL in FIG. 4.

Thus, the configuration of the shift stage included in the PMOS-typeshift register circuit has the connection relationship shown in FIG. 27.

The connection relationship among the respective thin film transistorsP21 to P26 is the same as that in the configuration of the NMOS-typeshift stage shown in FIG. 22.

The differences therebetween are that the higher potential Vdd isemployed as the first power supply to which one main electrode of eachof the thin film transistors P22, P23, and P24 is connected, and thatthe lower potential VSS is employed as the second power supply to whichone main electrode of each of the thin film transistors P25 and P26 isconnected.

FIG. 28 shows the drive waveforms of the shift stage according to thisform example. FIGS. 28A to 28F correspond to FIGS. 23A to 23F,respectively. The respective waveforms shown in FIG. 28 are obtained byinterchanging the H level and the L level of the respective waveformsshown in FIG. 23. That is, the basic operation is the same between theNMOS-type and PMOS-type shift stages. Therefore, the detaileddescription of the PMOS-type shift stage is omitted.

Also in the case of this PMOS-type shift stage, the load of the inputterminals in1 and in2 can be reduced compared with the related-artexample. As a result, the load of the transfer pulse is reduced. Thisallows driving at a higher frequency compared with the shift registercircuit according to the related-art example.

(B-2) SECOND FORM EXAMPLE OF SHIFT STAGE

In the above-described shift stage according to the first form example,the potential of the node B in the floating state depends on the leakagebalance between the thin film transistors N23 and N25 or P23 and P25.For example, in the NMOS-type shift stage, the leakages act to lower thepotential of the node B toward the lower potential VSS if the leakageamount of the thin film transistor N23 is larger than that of the thinfilm transistor N25. In contrast, the leakages act to increase thepotential of the node B toward the higher potential Vdd if the leakageamount of the thin film transistor N25 is larger than that of the thinfilm transistor N23.

In the PMOS-type shift stage, the potential of the node B is changed inthe direction opposite to that in the NMOS-type shift stage.

With reference to FIG. 29, a description will be made below about theinfluence when the leakage amount of the thin film transistor N25 islarger than that of the thin film transistor N23 in the NMOS-type shiftstage.

In this case, a possible issue is the drive operation in the periodduring which both the node A and the node B are in the floating state.

Originally, during this period, the node A is kept at the H level andthe node B is kept at the L level.

However, if the potential of the node B is shifted toward the H leveldue to the collapse of the leakage amount balance, the thin filmtransistor N24 is turned on when the potential of the node B reaches acertain potential. This acts to lower the potential of the node A in thefloating state toward the lower potential VSS, which possibly precludesthe bootstrap operation of the node A accompanying the input of theclock signal. Specifically, the amplitude of the pulse that appears atthe output node OUT is possibly insufficient.

To address this problem, in this form example, a circuit configurationis proposed in which the bootstrap operation of the node A is hardlyprecluded even with the collapse of the leakage amount balance due tothe influence of manufacturing variation.

It should be obvious that this form example is to further enhance thereliability and the shift stage according to the first form example canbe used without any problem as long as the influence of manufacturingvariation is within the allowable range.

(a) NMOS Type

FIG. 30 shows a second form example of the shift stage included in theshift register circuit. FIG. 31 shows the corresponding drive waveforms.

The difference between the shift stage shown in FIG. 30 and the shiftstage according to the first form example (FIG. 22) is the connectionform between the gate electrode of the thin film transistor N23 andanother node.

In this form example, the gate electrode of the thin film transistor N23is connected to the control interconnect of the thin film transistor N21(i.e. to the node A). The connection of the main electrodes of the thinfilm transistor N23 is the same as that in the first form example. Inthis form example, the thin film transistor N23 is turned on at the timeof the bootstrap operation of the node A to thereby act to fix thepotential of the node B at the lower potential VSS.

FIG. 31 shows the drive waveforms of the shift stage according to thisform example. The drive waveforms shown in FIGS. 31A to 31F correspondto those shown in FIGS. 23A to 23F, respectively.

As is apparent from comparison between FIG. 31E and FIG. 23E, in thisform example, the node B is not in the floating state but a fixedpotential is applied to the node B during the period when the node Acarries out bootstrap operation.

Specifically, even when the leakage amount of the thin film transistorN25 is relatively large, the potential of the node B can be continuouslykept at the lower potential VSS by the thin film transistor N23 that iscompletely turned on.

Thus, even if the degree of manufacturing variation is beyond theallowable range of the shift stage according to the first form example,employing the shift stage according to this form example makes itpossible to ensure normal bootstrap operation. That is, a shift registercircuit that carries out highly-reliable drive operation can beachieved.

(b) PMOS Type

The above-described problem relating to the leakage amount balanceapplies also to a PMOS-type shift stage.

Therefore, the same connection structure as that of the NMOS-type shiftstage can be employed for the PMOS-type shift stage.

FIG. 32 shows the circuit configuration of the PMOS-type shift stageaccording to this form example.

The difference between the shift stage shown in FIG. 32 and the shiftstage according to the first form example (FIG. 27) is also theconnection form of the gate electrode of the thin film transistor P23.

In this form example, the gate electrode of the thin film transistor P23is connected to the control interconnect of the thin film transistor P21(i.e. to the node A). This connection allows the thin film transistorP23 to fix the potential of the node B to the higher potential Vddduring the bootstrap operation of the node A.

Although the description of the drive waveforms of this PMOS-type shiftstage is omitted, employing the shift stage according to this formexample makes it possible to ensure normal bootstrap operation even ifthe degree of manufacturing variation is beyond the allowable range ofthe shift stage according to the first form example. That is, a shiftregister circuit that carries out highly-reliable drive operation can beachieved.

(B-3) THIRD FORM EXAMPLE OF SHIFT STAGE

The above-described shift stage according to the second form exampleinvolves a problem that, as shown in FIGS. 31D and 31E, the potentialswitching speed is low in switching of the potentials of the nodes A andB from the L level to the H level or from the H level to the L level.

This is because, even after the thin film transistor N25 (P25) or N26(P26) is turned on and thereby the potential supply is started, thepotential supply by the thin film transistor N23 (P23) or N24 (P24)continues for a while. That is, the potential supply by the first powersupply conflicts with that by the second power supply. In addition, theconfliction of the potential supply means that through-current flowsduring the confliction. This causes increase in the power consumption.Therefore, the time of the confliction of the power supply needs to beshortened as much as possible.

To meet this need, in this form example, a circuit configuration isproposed in which the bootstrap operation of the node A is hardlyprecluded even with the collapse of the leakage amount balance due tothe influence of manufacturing variation and fast switching of the nodepotentials can also be achieved. That is, a circuit configuration isproposed that carries out highly-reliable drive operation, and iscapable of high-frequency operation and has low power consumption.

(a) NMOS Type

FIG. 33 shows a third form example of the shift stage included in theshift register circuit.

The shift stage shown in FIG. 33 corresponds to a circuit configurationobtained by combining the shift stage according to the first formexample (FIG. 22) with the shift stage according to the second formexample (FIG. 30).

This circuit configuration arises from combination of the connectionform of the first form example, in which the speed of switching of thenode potentials is high, and the connection form of the second formexample, which is excellent in the potential holding performance (latchperformance).

In this form example, the thin film transistor N23 corresponding to thefirst form example is indicated as N23(1), and the thin film transistorN23 corresponding to the second form example is indicated as N23(2).Similarly, the thin film transistor N24 corresponding to the first formexample, in which the speed of node potential switching is high, isindicated as N24(1), and the thin film transistor N24 that correspondsto the first and second form examples and is excellent in the potentialholding performance (latch performance) is indicated as N24(2).

In this form example, when the H-level potential is input to the firstinput terminal in1, the thin film transistor N23(1) is turned on first,so that the lowering of the potential of the node B, which has been inthe floating state so far, is immediately started. Thereafter, the thinfilm transistor N23(2) is also turned on, and the potential of the nodeB is completely lowered to the lower potential VSS.

Because the potential of the node B is lowered fast in this manner, thethin film transistor N24(2) is also rapidly turned off to stop thesupply of the lower potential VSS to the node A. Thus, the speed of thepotential rise of the node A to the higher potential Vdd is alsoimproved.

Furthermore, when the H-level potential is input to the second inputterminal in2, the thin film transistor N24(1) is turned on first, sothat the lowering of the potential of the node A, which has been in thefloating state so far, is immediately started. Thereafter, the thin filmtransistor N24(2) is also turned on, and the potential of the node A iscompletely lowered to the lower potential VSS.

Because the potential of the node A is lowered fast in this manner, thethin film transistor N23(2) is also rapidly turned off to stop thesupply of the lower potential VSS to the node B. Thus, the speed of thepotential rise of the node B to the higher potential Vdd is alsoimproved.

FIG. 34 shows the drive waveforms of the shift stage according to thisform example. The drive waveforms shown in FIGS. 34A to 34F correspondto those shown in FIGS. 31A to 31F, respectively.

As is apparent from comparison between FIGS. 34D and 34E and FIGS. 31Dand 31E, the speed of the potential change of the nodes A and B ishigher in this form example.

In addition, as is apparent from comparison with FIG. 23E, the supply ofthe lower potential VSS to the node B is continued also during thebootstrap operation of the node A, due to the provision of the thin filmtransistor N23(2).

Therefore, in this form example, the normal bootstrap operation of thenode A can be ensured even if the leakage amount of the thin filmtransistor N25 is relatively large due to the influence of manufacturingvariation.

That is, using the shift stage according to this third form examplemakes it possible to achieve a shift register circuit in which the speedof switching of the node potentials is high and the reliability of thedrive operation is high. In addition, increase in the power consumptioncan also be minimized in spite of the enhancement in the potentialswitching speed.

(b) PMOS Type

The above-described problem applies also to a PMOS-type shift stage.

Therefore, the same connection structure as that of the NMOS-type shiftstage can be employed for the PMOS-type shift stage.

Specifically, a circuit configuration obtained by combining the shiftstage according to the first form example (FIG. 27) with the shift stageaccording to the second form example (FIG. 32) is employed.

FIG. 35 shows the circuit configuration of the PMOS-type shift stageaccording to this form example.

In the shift stage shown in FIG. 35, the thin film transistor P23corresponding to the first form example is indicated as P23(1), and thethin film transistor P23 corresponding to the second form example isindicated as P23(2). Similarly, the thin film transistor P24corresponding to the first form example, in which the speed of nodepotential switching is high, is indicated as P24(1), and the thin filmtransistor P24 that corresponds to the first and second form examplesand is excellent in the potential holding performance (latchperformance) is indicated as P24(2).

Although the description of the drive waveforms of this PMOS-type shiftstage is omitted, employing the shift stage according to this formexample makes it possible to achieve a shift register circuit having thesame advantages as those of the shift register circuit including theNMOS-type shift stage.

(B-4) FOURTH FORM EXAMPLE OF SHIFT STAGE

As described above, using the shift stage according to the third formexample can prevent the bootstrap operation of the node A from beingaffected by the influence of production variation.

However, the period during which the node B is kept at the floatingstate at the H level involves a slight possibility that off-leakage dueto production variation leads to erroneous output to the output nodeOUT.

FIG. 36 shows leakage paths that possibly become an issue in the shiftstage according to the third form example, regarding the NMOS-type shiftstage. The problematic leakage paths are the leakage path through thethin film transistor N25 and two leakage paths through the thin filmtransistors N23(1) and N23(2).

If the sum of the leakage amounts of the thin film transistors N23(1)and N23(2) is larger than the leakage amount of the thin film transistorN25, the potential of the node B, which should be kept at the higherpotential Vdd originally, is lowered toward the lower potential VSS.

The potential decrease of the node B in the period of the floating statethereof causes the occurrence of the floating state of the node A asdescribed above with FIG. 14. This floating state of the node A causes aphenomenon that the clock signal input from the clock input terminal CKvia the coupling capacitance turns on the thin film transistor N21 andan erroneous transfer pulse appears at the output node OUT.

To address this problem, in this form example, it is proposed to use aconfiguration in which the leakage amount of the thin film transistorN25 is structurally larger than that of the thin film transistors N23(1)and N23(2), with focus on the characteristic that the second powersupply Vdd can be utilized. That is, a method is proposed in which theoff-leakage that causes the potential of the node B to be lowered towardthe lower potential VSS is structurally decreased to thereby enhance thereliability of the shift operation.

(a) Method of Utilizing Difference in Transistor Size

As a first method, the present inventors focus attention on thetransistor size. Specifically, the present inventors focus attention onthe characteristic that the off-leakage through a thin film transistorbecomes larger as the ratio of the channel width W of the thin filmtransistor to the channel length L thereof (i.e. W/L) becomes higher.

Based on this characteristic, the transistor sizes are so determinedthat W/L of the thin film transistor N25 is larger than W/L of the thinfilm transistors N23(1) and N23(2).

FIG. 37 shows a configuration example of the NMOS-type shift stage, andFIG. 38 shows a configuration example of the PMOS-type shift stage.

As long as the sizes of the thin film transistors surrounded by thedashed line and the dotted line in the diagrams satisfy theabove-described relationship, the other thin film transistors may haveany size.

Therefore, the other thin film transistors may have the same size asthat of the thin film transistors N23(1) and N23(2) or P23(1) andP23(2), or may have the same size as that of the thin film transistorN25 or P25. Alternatively, they may have a third size different fromthese sizes.

In any case, this larger off-leakage of the thin film transistor N25 orP25 makes it easy to hold the potential of the node B that is at the Hlevel and in the floating state. This can enhance the reliability of thedrive operation.

(b) Method of Utilizing Difference in Gate Structure of Transistor

As a second method, the present inventors focus attention on the gatestructure. FIG. 39 shows the sectional structures of thin filmtransistors with focus on difference in the gate structure.

FIGS. 39A and 39B show thin film transistors having a structure in whicha gate electrode is formed on merely one side of the channel layer.Hereinafter, this structure will be referred to as the single-sided gatestructure.

The structure shown in FIG. 39A, in which the gate electrode is disposedunder the channel layer, is referred to as the bottom-gate structure.

The structure shown in FIG. 39B, in which the gate electrode is disposedover the channel layer, is referred to as the top-gate structure.

FIG. 39C shows a thin film transistor having a structure in which gateelectrodes are formed on both the sides of the channel layer (over andunder the channel layer). Hereinafter, this structure will be referredto as the double-sided gate structure. In the structure of FIG. 39C, theupper gate electrode and the lower gate electrode are controlled by thesame power supply. However, it is also possible to control them bydifferent power supplies. Using different power supplies can shift theoperating point, and thus can control the off-leakage.

It is known that the intensity of the gate-source electric field in thethin film transistor having the double-sided gate structure is higherthan that of the gate-source electric field in the thin film transistorhaving the single-sided gate structure. Therefore, it is generally knownthat the thin film transistor with the double-sided gate structure has acharacteristic of having large on-current and a small S value.

FIG. 40 shows the result of actual measurement relating to therelationship between the gate structure and the off-leakage. FIG. 40Ashows the relationship between the gate-source voltage Vgs and thedrain-source current Ids in an NMOS thin film transistor. FIG. 40B showsthe relationship between the gate-source voltage Vgs and thedrain-source current Ids in a PMOS thin film transistor.

To show the off-leakage current, the ordinates in FIGS. 40A and 40B arerepresented on the log scale.

Comparison at the operating point of the gate-source voltage Vgs=0 Vmakes it apparent that the off-leakage current of the thin filmtransistor having the double-sided gate structure is smaller than thatof the thin film transistor having the single-sided gate structure bytwo to three orders of magnitude. This means that the off-resistance ofthe single-sided gate structure is lower than that of the double-sidedgate structure by two to three orders of magnitude.

Based on this characteristic, in this method, the single-sided gatestructure is employed for the thin film transistor N25 or P25, and thedouble-sided gate structure is employed for the thin film transistorsN23(1) and N23(2) or P23(1) and P23(2).

FIG. 41 shows a configuration example of the NMOS-type shift stage, andFIG. 42 shows a configuration example of the PMOS-type shift stage.

As long as the gate structures of the thin film transistors surroundedby the dashed line and the dotted line in the diagrams satisfy theabove-described relationship, the other thin film transistors may haveany gate structure.

Therefore, the gate structure of each of the other thin film transistorsmay be either the single-sided gate structure or the double-sided gatestructure.

In any case, this larger off-leakage of the thin film transistor N25 orP25 makes it easy to hold the potential of the node B that is at the Hlevel and in the floating state. This can enhance the reliability of thedrive operation.

(c) Method of Utilizing Difference in Source-Shield Length of Transistor

As a third method, the present inventors focus attention on thesource-shield length. Initially, the relationship between thesource-shield length and the off-leakage amount will be described belowwith reference to FIGS. 43 and 44.

FIG. 43A shows the sectional structure of an NMOS thin film transistor.

In the case of the NMOS thin film transistor, the source-shield lengthrefers to the length of the partial portion of the metal interconnect(source electrode) connected to the source, as the part covering thechannel region. A lightly doped drain (LDD) region is not included inthis channel region.

FIG. 44A shows the sectional structure of a PMOS thin film transistor.Also in the case of the PMOS thin film transistor, the source-shieldlength refers to the length of the partial portion of the metalinterconnect (source electrode) connected to the source, as the partcovering the channel region. In the structure of FIG. 44A, the PMOS thinfilm transistor does not have an LDD layer.

FIG. 43B shows the relationship between the gate-source voltage Vgs andthe drain-source current Ids in the NMOS thin film transistor. FIG. 44Bshows the relationship between the gate-source voltage Vgs and thedrain-source current Ids in the PMOS thin film transistor. In eithercase, the ordinate is represented on the log scale.

As shown in FIGS. 43B and 44B, the following characteristic is found ineither case: smaller source-shield length provides larger off-leakageand larger source-shield length provides smaller off-leakage.

Based on this characteristic, in this method, a thin film transistorhaving a relatively-small source-shield length is employed as the thinfilm transistor N25 or P25, and thin film transistors having arelatively-large source-shield length are employed as the thin filmtransistors N23(1) and N23(2) or P23(1) and P23(2).

FIG. 45 shows a configuration example of the NMOS-type shift stage, andFIG. 46 shows a configuration example of the PMOS-type shift stage.

As long as the source-shield lengths of the thin film transistorssurrounded by the dashed line and the dotted line in the diagramssatisfy the above-described relationship, the other thin filmtransistors may have any source-shield length.

Therefore, the other thin film transistors may have a source-shieldlength larger than the above-described two kinds of source-shieldlengths, or one between the above-described two kinds of source-shieldlengths, or one smaller than the above-described two kinds ofsource-shield lengths.

In any case, this larger off-leakage of the thin film transistor N25 orP25 makes it easy to hold the potential of the node B that is at the Hlevel and in the floating state. This can enhance the reliability of thedrive operation.

The purpose of the provision of the LDD region in the NMOS thin filmtransistor is to greatly reduce the off-leakage by electric fieldrelaxation.

If the NMOS thin film transistor does not have the LDD region, theoff-leakage current is on the order of 10̂-8. In contrast, if it has theLDD region, the off-leakage current can be decreased to one on the orderof 10̂-13, although depending on the LDD length. On the other hand, theoff-leakage current in the PMOS thin film transistor is on the order of10̂-13 without the LDD region. Therefore, there is no need to apply theLDD structure, which increases the number of processes, to the PMOS thinfilm transistor. The desired value of the off-leakage current in drivingis 10̂-10 or smaller.

If the source and the drain are interchanged (the metal interconnect isconnected to the drain) in the structures of FIGS. 43 and 44, thecharacteristic can be shifted depending on the drain shield. As theshield length of the drain shield is increased, the TFT characteristicis further shifted toward the depletion side. Thus, at the circuitoperating point of Vgs=0, larger shield length provides largeroff-leakage.

However, the amount of the TFT characteristic shift in relation to thedrain shield depends greatly on the drain potential. In addition, thedrain potential varies and thus is difficult to control in the operation(for a TFT, the source potential is the reference potential and thedrain potential is regarded as the relative potential difference withrespect to the source potential). Although it is possible to control theoff-leakage by using the drain shield in principle, the off-leakage islarge at the operating point in the form examples of the presentinvention and thus erroneous operation possibly occurs. Therefore, it isdifficult to utilize the drain shield.

In the case of the source shield, larger shield length shifts the TFTcharacteristic toward the enhancement side with a higher degree, andlarger shield length provides smaller off-leakage at the circuitoperating point of Vgs=0. Because the source potential is the referencepotential for a TFT, the source shield gives the electrically-stable TFTcharacteristic.

(d) Method of Utilizing Difference in LDD Length of Transistor

The above-described three methods can be applied to both NMOS and PMOStransistors. However, another method other than these methods is alsoavailable as long as this method is applied to an NMOS thin filmtransistor.

In this method as a fourth method, the present inventors focus attentionon the LDD length of an NMOS thin film transistor. The LDD region refersto the low-concentration impurity region provided between thesource/drain and the channel as shown in FIG. 43A. The LDD region isused to avoid focusing of a high electric field on the correspondingregion part. It is known that smaller LDD length provides largeroff-leakage.

Based on this characteristic, in this method, a thin film transistorhaving a relatively-small LDD length is employed as the thin filmtransistor N25, and thin film transistors having a relatively-large LDDlength are employed as the thin film transistors N23(1) and N23(2).

FIG. 47 shows a configuration example of the NMOS-type shift stage.

As long as the LDD lengths of the thin film transistors surrounded bythe dashed line and the dotted line in the diagram satisfy theabove-described relationship, the other thin film transistors may haveany LDD length.

Therefore, the other thin film transistors may have an LDD length largerthan the above-described two kinds of LDD lengths, or one between theabove-described two kinds of LDD lengths, or one smaller than theabove-described two kinds of LDD lengths.

In any case, this larger off-leakage of the thin film transistor N25makes it easy to hold the potential of the node B that is at the H leveland in the floating state. This can enhance the reliability of the driveoperation.

(B-5) FIFTH FORM EXAMPLE OF SHIFT STAGE

In the following, a technique to decrease the potential amplitude of thefirst and second input terminals in1 and in2 will be described below.Specifically, a description will be made below about the configurationof a shift stage in which amplitude of VSS/Vdd appears at the outputnode OUT even if the amplitude of the input to the first and secondinput terminals in1 and in2 is not VSS/Vdd.

However, the shift stages according to the above-described respectiveform examples are premised on the shift register circuit having theconnection structure shown in FIG. 21. Therefore, to the first andsecond input terminals of the shift stages other than the first shiftstage and the final shift stage, the transfer pulse with the maximumamplitude of VSS/Vdd is input. As a result, the pulses whose amplitudecan be decreased are limited to two pulses: the start pulse st input tothe first shift stage; and the end pulse ‘end’ input to the final shiftstage.

However, the amplitude decrease of these two pulses means that anamplitude decrease for the generation circuits of these pulses can beachieved. If the amplitude is small, a voltage decrease of the drivepower supply desired for the respective generation circuits of these twopulses can also be achieved. Thus, reduction in the power consumption ofthese two pulses can be achieved.

(a) NMOS Type

FIG. 48 shows a fifth form example of the shift stage included in theshift register circuit. FIG. 49 shows the corresponding drive waveforms.The drive waveforms shown in FIGS. 49A to 49F correspond to those shownin FIGS. 31A to 31F, respectively.

The configuration of the shift stage shown in FIG. 48 is the same asthat of the shift stage according to the third form example (FIG. 33).

However, as shown in FIGS. 49A and 49B, the H-level potential of thepulse signal input to the first and second input terminals in1 and in2is Vin lower than the potential Vdd of the first power supply.

Therefore, the operation condition desired for the shift stage shown inFIG. 48 is that the transfer pulse with the maximum amplitude appears atthe output node OUT due to the bootstrap operation of the node A evenwith the input of the pulse signal having the smaller amplitude.

FIG. 50 shows the drive condition desired for the node A. In FIG. 50,the lower potential VSS is defined as 0 V.

The potential of the node A except for in the bootstrap operationthereof is represented as Vin−Vth(N26) if the higher potential Vin isinput to the first input terminal in1.

In this case, the amplitude at the time of the bootstrap operation isrepresented as {(C1+C(N21))/(C1+C(N21)+Cpa)}*Vdd.

Therefore, if the value obtained by subtracting the output potential Vddfrom the maximum potential, which arises from addition of the amplitudepotential at the time of the bootstrap operation to the potentialVin−Vth(N26) obtained before the bootstrap operation, is equal to orhigher than the threshold voltage Vth(N21) of the thin film transistorN21, the same operation as that in the above-described respective formexamples can be achieved although the amplitude of the start pulse stand the end pulse ‘end’ is decreased.

C(N21) denotes the gate capacitance of the thin film transistor N21, andCpa denotes the total parasitic capacitance of the node A, except forC(N21) and C1.

At the time of the design, the values of the complementary capacitanceC1, the gate capacitance C(N21), and the total parasitic capacitance Cpaare optimized so that the above-described condition may be satisfied.

FIG. 51 shows drive pulse waveforms of the shift register circuitincluding the shift stage according to this form example. The drivewaveforms shown in FIGS. 51A to 51G correspond to those shown in FIGS.18A to 18G, respectively.

It can be understood from FIG. 51 that the output pulse having the sameamplitude as that of the clock signal appears at the output node of eachshift stage although the signal amplitude of the start pulse st and theend pulse ‘end’ is smaller as shown in FIGS. 51C and 51G.

Although the third form example is taken as an example in the abovedescription, the technique to decrease the amplitude of the start pulsest and the end pulse ‘end’ can be applied also to the first, second, andfourth form examples similarly.

(b) PMOS Type

The above-described technique to decrease the amplitude of the startpulse st and the end pulse ‘end’ can be applied also to a PMOS-typeshift stage in the same manner. The desired parameter condition is thesame as that of the NMOS-type shift stage.

FIG. 52 shows the circuit configuration of the PMOS-type shift stageaccording to this form example. Although the description of the drivewaveforms of this PMOS-type shift stage is omitted, employing the shiftstage according to this form example makes it possible to achieve ashift register circuit having the same advantages as those of the shiftregister circuit including the NMOS-type shift stage.

(B-6) SIXTH FORM EXAMPLE OF SHIFT STAGE

A description will be made below about specific structure examples toachieve the amplitude decrease of the start pulse st and the end pulse‘end’ input to the first and second input terminals in1 and in2 in thefifth form example.

(a) NMOS Type

In order to achieve the amplitude decrease of the pulse signal, astructure having a high bootstrap gain Gbt, represented by the followingequation, should be employed.

Gbt=(C1+C(N21))/(C1+C(N21)+Cpa)

To increase the bootstrap gain Gbt, the total parasitic capacitance Cpaof the node A should be decreased. Alternatively, the complementarycapacitance C1 and the gate capacitance C(N21) of the thin filmtransistor N21 should be increased.

In the present example, attention is focused on the gate capacitanceC(N21) of the thin film transistor N21.

The relationship between the structure of the thin film transistor andthe gate capacitance C(N21) will be described below with reference toFIG. 53.

If the gate capacitance of the thin film transistor having thebottom-gate structure shown in FIG. 53A is defined as Ca and the gatecapacitance of the thin film transistor having the top-gate structureshown in FIG. 53B is defined as Cb, the gate capacitance of the thinfilm transistor having a double-sided gate structure is represented asCa+Cb. That is, by providing the thin film transistor N21 with thedouble-sided gate structure, the bootstrap gain can be increased. Higherbootstrap gain provides higher operation stability of the shift stage.

FIG. 54 shows a sixth form example of the shift stage included in theshift register circuit. As shown in FIG. 54, the thin film transistorN21 having a double-sided gate structure is employed in the shift stageaccording to this form example. The double-sided gate structureintensifies the gate-source electric field. Thus, the on-current isincreased. Therefore, the double-sided gate structure provides anadvantage of enhancing the speed of the rising and falling (so-calledtransient) of the current amount of the output node OUT as shown in FIG.55.

FIG. 55A shows the current characteristic of an NMOS thin filmtransistor, and FIG. 55B shows the current characteristic of a PMOS thinfilm transistor. In either case, the double-sided gate structureprovides sharper change in the current amount.

(b) PMOS Type

The above-described technique to increase the bootstrap gain can beapplied also to a PMOS-type shift stage in the same manner. Therefore,the circuit configuration shown in FIG. 56 can be employed for thePMOS-type shift stage according to this form example. Although thedescription of the drive waveforms of this PMOS-type shift stage isomitted, employing the shift stage according to this form example makesit possible to simultaneously achieve lower power consumption andhigher-frequency driving as with the NMOS-type shift stage.

(B-7) SEVENTH FORM EXAMPLE OF SHIFT STAGE

The shift stages according to the above-described first to sixth formexamples are premised on the connection form shown in FIG. 21.Specifically, in these form examples, the transfer pulse that appears atthe output node OUT of each shift stage is coupled not merely to thecontrol line corresponding to this shift stage but also to the secondinput terminal in2 of the previous shift stage and the first inputterminal in1 of the subsequent shift stage.

In the following, a shift register circuit having an interconnectstructure smaller than that of FIG. 21 will be described below. In thisform example, a third clock signal ck3 is added instead of the end pulse‘end’ and is used for driving of the clock input terminal CK and thesecond input terminal in2. In addition, in linkage with this, theconnection between the output node of the shift stage and the secondinput terminal of the previous shift stage is eliminated.

FIG. 57 shows the interconnect structure of the shift register circuitproposed in this form example. As shown in FIG. 57, the output node OUTof the shift stage is connected merely to the control line correspondingto this shift stage and the first input terminal in1 of the subsequentshift stage. On the other hand, of three clock signals CKA, CKB, and CKCwhose phases are shifted from each other by a 1H period, two clocksignals whose phases are adjacent to each other are coupled to the clockinput terminal CK and the second input terminal in2 of each shift stage,with the cyclically-shifted phase relationship.

In the structure of FIG. 57, the first clock signal CKA is input to theclock input terminal CK of the first shift stage, and the second clocksignal CKB is input to the second input terminal in2 of the first shiftstage. The second clock signal CKB is input to the clock input terminalCK of the second shift stage, and the third clock signal CKC is input tothe second input terminal in2 of the second shift stage. The third clocksignal CKC is input to the clock input terminal CK of the third shiftstage, and the first clock signal CKA is input to the second inputterminal in2 of the third shift stage. This connection structure isrepeated also for the subsequent shift stages. That is, the phases ofthe clock signals input to the clock input terminal CK and the secondinput terminal in2 are cyclically shifted on a stage-by-stage basis.

(a) NMOS Type

FIG. 58 shows an example of drive pulse waveforms of the shift registercircuit formed with merely NMOS thin film transistors.

FIG. 58A shows the first clock signal CKA. FIG. 58B shows the secondclock signal CKB. FIG. 58C shows the third clock signal CKC.

FIG. 58D shows the start pulse st for driving the first shift stage.FIG. 58E shows an output pulse o1 of the first shift stage SR(1). FIG.58F shows an output pulse o2 of the second shift stage SR(2). FIG. 58Gshows an output pulse o3 of the third shift stage SR(3).

As shown in FIG. 58, from the (3m+1)-th shift stage (m is 0, 1, 2, . . .), the output pulse is output at the timing of the first clock signalCKA. From the (3m+2)-th shift stage (m is 0, 1, 2, . . . ), the outputpulse is output at the timing of the second clock signal CKB. From the(3m+3)-th shift stage (m is 0, 1, 2, . . . ), the output pulse is outputat the timing of the third clock signal CKC.

It is obvious that the input of the output pulse from the previous shiftstage to the first input terminal in1 is necessary for the output of theoutput pulse. Therefore, the output pulse is transferred in such amanner as to be shifted on a stage-by-stage basis sequentially from thefirst shift stage.

FIG. 59 shows the drive waveforms corresponding to each shift stage. Thedrive waveforms of FIGS. 59A to 59F correspond to those of FIGS. 34A to34F, respectively.

The largest difference in the drive waveforms between FIGS. 59 and 34 isthat the clock pulse is periodically input (specifically, with a3H-cycle) to the second input terminal in2 as shown in FIG. 59B.

This clock pulse input means that the H-level potential (higherpotential Vdd) can be periodically supplied to the node B. Therefore, inthe shift register circuit according to this form example, an advantagethat the floating-state period of the node B can be shortened can beexpected.

The shortening of the floating-state period of the node B means that thepotentials of the nodes A and B in the period other than the outputtiming of the output pulse can be so kept as to have the normalpotential relationship. Thus, this is effective for enhancement in thereliability of the drive operation, such as prevention of the output ofthe output pulse at an erroneous timing.

(b) PMOS Type

Also for a PMOS-type shift register circuit, the same connection form asthat of the NMOS-type shift register circuit can be employed. Thepotential relationship of the drive waveforms is opposite to that of theNMOS-type shift register circuit, similarly to the other form examples.

FIG. 60 shows an example of drive pulse waveforms of a shift registercircuit formed with merely PMOS thin film transistors.

FIG. 60A shows the first clock signal CKA. FIG. 60B shows the secondclock signal CKB. FIG. 60C shows the third clock signal CKC.

FIG. 60D shows the start pulse st for driving the first shift stage.

FIG. 60E shows an output pulse o1 of the first shift stage SR(1). FIG.60F shows an output pulse o2 of the second shift stage SR(2). FIG. 60Gshows an output pulse o3 of the third shift stage SR(3).

Also in this case, as shown in FIG. 60, from the (3m+1)-th shift stage(m is 0, 1, 2, . . . ), the output pulse is output at the timing of thefirst clock signal CKA. From the (3m+2)-th shift stage (m is 0, 1, 2, .. . ), the output pulse is output at the timing of the second clocksignal CKB. From the (3m+3)-th shift stage (m is 0, 1, 2, . . . ), theoutput pulse is output at the timing of the third clock signal CKC.

It is obvious that the input of the output pulse from the previous shiftstage to the first input terminal in1 is necessary for the output of theoutput pulse. Therefore, the output pulse is transferred in such amanner as to be shifted on a stage-by-stage basis sequentially from thefirst shift stage.

FIG. 61 shows the drive waveforms corresponding to each shift stage. Thedrive waveforms of FIGS. 61A to 61F correspond to those of FIGS. 28A to28F, respectively.

The largest difference in the drive waveforms between FIGS. 61 and 28 isthat the clock pulse is periodically input (specifically, with a3H-cycle) to the second input terminal in2 as shown in FIG. 61B.

This clock pulse input means that the L-level potential (lower potentialVSS) can be periodically supplied to the node B. Therefore, in the shiftregister circuit according to this form example, an advantage that thefloating-state period of the node B can be shortened can be expected.

The shortening of the floating-state period of the node B means that thepotentials of the nodes A and B in the period other than the outputtiming of the output pulse can be so kept as to have the normalpotential relationship. Thus, this is effective for enhancement in thereliability of the drive operation, such as prevention of the output ofthe output pulse at an erroneous timing.

(B-8) EIGHTH FORM EXAMPLE OF SHIFT STAGE

In each of the shift stages according to the above-described first toseventh form examples, as shown in FIG. 62, the first input terminal in1is connected to the gate electrode of the thin film transistor N26, andthe second input terminal in2 is connected to the gate electrode of thethin film transistor N25.

Due to this connection structure, as shown in FIG. 63, the potentials ofthe node A and the node B are lower than the higher potential Vdd by thethreshold voltages Vth(N25) and Vth(N26) of the thin film transistorsN25 and N26.

The cause of this state is that the thin film transistors N25 and N26become the diode-connected state when the higher potential Vdd isapplied to the first and second input terminals in1 and in2.

FIG. 64 shows the equivalent circuit of the thin film transistor N26when the higher potential Vdd is applied. Via this diode-connected thinfilm transistor N26, the node A is charged from the lower potential VSStoward the higher potential Vdd. However, as shown in FIG. 65, thecharging of the node A progresses merely to such a degree that therelationship Vgs=Vds=Vth is obtained.

The reason for this is that the diode-connected thin film transistor N26operates in the saturation region.

In the diode connection of FIG. 64, the source region of the thin filmtransistor N26 is connected to the node A. Therefore, the sourcepotential Vs also rises up along with the charging of the node A.Specifically, the charging of the node A (the increase in the sourcepotential Vs) acts to decrease the drain-source voltage Vds (i.e. Vgs).

The decrease in the drain-source voltage Vds acts to decrease the draincurrent Ids as shown in FIG. 66. Specifically, as the potential of thenode A increases, the on-resistance of the thin film transistor N26 alsoincreases. In due course, the thin film transistor N26 is turned off atthe timing when the relationship Vgs=Vds=Vth(N26) is obtained. This isthe reason why the node A may not be charged to the higher potentialVdd.

The operation of the thin film transistors N25 and N26 as thediode-connected elements is carried out at the timing of the switchingbetween periods t1 and t2 in FIG. 63 (i.e. the timing of the switchingof the potential of the node A) and at the timing of the switchingbetween periods t5 and t6 (i.e. the timing of the switching of thepotential of the node B).

The speed of the rising of the node potential at these timings dependson the on-resistance of the thin film transistors N25 and N26 and theparasitic capacitance of the node.

As described above, the on-resistance of the thin film transistors N25and N26 increases along with potential increase of the nodes A and B.Therefore, the speed of the rising of the node potential becomes lower.

The switching between the periods t1 and t2 in FIG. 63 (i.e. theswitching of the potential of the node A) is preparatory operation forbootstrap operation. Therefore, it is desired to decrease theon-resistance of the thin film transistor N21 as much as possible. Thus,it is the best to surely cut off the thin film transistor N26.

On the other hand, the switching between the periods t5 and t6 in FIG.63 (i.e. the switching of the potential of the node B) is to end theshift operation, and the potential of the node A needs to have fallendown by the time the clock signal rises up to the H level next time. Ifthe potential of the node A has not fallen down, erroneous transferoccurs.

That is, the transient of this potential switching (rising speed)defines the drive frequency. Enhancing the speed of this potentialswitching allows driving at a higher frequency.

The function of the thin film transistors N23(2) and N24(2) for fixingthe potentials of the nodes A and B at the lower potential VSS is mostlysuppressed by the thin film transistors N23(1) and N24(1).

(a) NMOS Type

FIG. 67 shows an eighth form example of a shift stage suitable forhigh-frequency driving. FIG. 68 shows the corresponding drive waveforms.

The same parts in FIG. 67 as those in the shift stages according to theabove-described respective form examples are given the same symbols.

As the configuration specific to this form example, thin filmtransistors N27 and N28 and complementary capacitors C3 and C4 are addedso that the gate potentials of the thin film transistors N25 and N26 maybe raised by bootstrap operation. That is, this form example is definedin that the thin film transistors N25 and N26 can be utilized in thelinear region.

Specifically, the gate electrode of the thin film transistor N25 isconnected to the second input terminal in2 via the thin film transistorN27. Furthermore, the complementary capacitor C3 for complementingbootstrap operation is connected between the gate electrode of the thinfilm transistor N25 and the main electrode thereof closer to the node B.

Naturally, the complementary capacitor C3 is unnecessary if the gatecapacitor of the thin film transistor N25 has sufficiently-highcapacitance for the bootstrap operation.

The gate electrode of the newly-added thin film transistor N27 isconnected to the second power supply (Vdd). Due to this connection, thethin film transistor N27 is typically kept at the on-state. Therefore,the power is typically supplied to the gate electrode of the thin filmtransistor N25 (i.e. to a node D).

Similarly, in this form example, the gate electrode of the thin filmtransistor N26 is connected to the first input terminal in1 via the thinfilm transistor N28. Furthermore, the complementary capacitor C4 forcomplementing bootstrap operation is connected between the gateelectrode of the thin film transistor N26 and the main electrode thereofcloser to the node A.

Naturally, the complementary capacitor C4 is unnecessary if the gatecapacitor of the thin film transistor N26 has sufficiently-highcapacitance for the bootstrap operation.

The gate electrode of the newly-added thin film transistor N28 is alsoconnected to the second power supply (Vdd). Due to this connection, thethin film transistor N28 is typically kept at the on-state. Therefore,the power is typically supplied to the gate electrode of the thin filmtransistor N26 (i.e. to a node C).

FIG. 68 shows the drive waveforms of the shift stage according to thisform example. FIG. 68A shows the input waveform of the first inputterminal in1. FIG. 68B shows the input waveform of the second inputterminal in2. FIG. 68C shows the input waveform of the clock signal ck.FIG. 68D shows the drive waveform of the node C. FIG. 68E shows thedrive waveform of the node D. FIG. 68F shows the drive waveform of thenode A. FIG. 68G shows the drive waveform of the node B. FIG. 68H showsthe signal waveform of the output pulse that appears at the output nodeOUT.

The basic drive timings and the operation of potential switching at thenodes A and B in this form example are the same as those in the otherform examples. Specifically, as in the other form examples, thepotential of the node A rises up to the H level at the timing when theH-level potential (Vdd) is input to the first input terminal in1, andthe potential of the node A falls down to the L level at the timing whenthe H-level potential (Vdd) is input to the second input terminal in2.

In the following, the operation of the thin film transistors N25 and 26,which is specific to this form example, will be mainly described below.

As described above, the addition of the thin film transistors N27 andN28 allows the gate potentials of the thin film transistors N25 and N26to be raised by bootstrap operation.

This makes it possible to utilize the thin film transistors N25 and N26in the linear region. In this case, the thin film transistors N25 andN26 can be equivalently regarded as a resistive element as shown inFIGS. 69A and 69B. The resistance of this resistive element depends onthe H-level potential of the nodes C and D obtained due to the bootstrapoperation.

Therefore, by setting the H-level potential of the node C (FIG. 68D)equal to or higher than the potential obtained by adding the thresholdvoltage Vth(N26) of the thin film transistor N26 to Vdd, the H-levelpotential of the node A can be raised to Vdd as shown in FIG. 70.

Similarly, by setting the H-level potential of the node D (FIG. 68E)equal to or higher than the potential obtained by adding the thresholdvoltage Vth(N25) of the thin film transistor N25 to Vdd, the H-levelpotential of the node B can be raised to Vdd as shown in FIG. 70.

The equivalent resistance of the thin film transistors N25 and N26depends on the relationship between the H-level potential of the nodes Cand D and the higher potential Vdd. Thus, the thin film transistors N25and N26 are not turned off even when the potential of the nodes A and Bis increased.

As above, in the shift stage according to this form example, the nodes Aand B can be charged to the higher potential Vdd rapidly due to thebootstrap operation of the nodes C and D. Therefore, this form exampleis advantageous in high-frequency operation.

Furthermore, the potential of the node A (FIG. 68F) in the periods t2and t3 in FIG. 68 is set to Vdd (maximum potential). That is, thepotential of the node A before the bootstrap operation thereof can beset higher than that in the above-described other form examples.

Thus, the bootstrap amount desired for the thin film transistor N21 canbe lowered. As a result, size reduction or elimination of thecomplementary capacitor C1 can be achieved. This allows the layout areato be decreased compared with the shift stages according to the otherform examples.

Moreover, in this form example, the H-level potential held by the node B(FIG. 68G) can also be raised to Vdd. Thus, the margin of the leakagetime can be widened.

The above-described features allow the shift stage according to thisform example to offer a shift stage and a shift register circuit thatare excellent in high-speed operation and operation stability and havesmall layout area.

It is also possible to use the configuration of this shift stage incombination with the configuration of any of the shift stages accordingto the above-described respective form examples.

(b) PMOS Type

Also for a PMOS-type shift register circuit, the same connection form asthat of the NMOS-type shift register circuit can be employed. Thepotential relationship of the drive waveforms is opposite to that of theNMOS-type shift register circuit, similarly to the other form examples.

FIG. 71 shows the eighth form example of a shift stage suitable forhigh-frequency driving. FIG. 72 shows the corresponding drive waveforms.

As shown in FIG. 71, in the PMOS-type shift stage, thin film transistorsP27 and P28 and the complementary capacitors C3 and C4 are newly addedso that the gate potentials of the thin film transistors P25 and P26 maybe raised by bootstrap operation. This feature allows the thin filmtransistors P25 and P26 to operate in the linear region.

The details of the drive waveforms shown in FIG. 72 are as follows. FIG.72A shows the input waveform of the first input terminal in1. FIG. 72Bshows the input waveform of the second input terminal in2. FIG. 72Cshows the input waveform of the clock signal ck. FIG. 72D shows thedrive waveform of the node C. FIG. 72E shows the drive waveform of thenode D. FIG. 72F shows the drive waveform of the node A. FIG. 72G showsthe drive waveform of the node B. FIG. 72H shows the signal waveform ofthe output pulse that appears at the output node OUT.

As shown in FIG. 72, in the shift stage according to this form example,the nodes A and B can be discharged to the lower potential VSS rapidlydue to the bootstrap operation of the nodes C and D. Therefore, thisform example is advantageous in high-frequency operation.

Furthermore, the potential of the node A (FIG. 72F) in the periods t2and t3 in FIG. 72 is set to VSS (minimum potential). That is, thepotential of the node A before the bootstrap operation thereof can beset lower than that in the above-described other form examples.

Thus, the bootstrap amount desired for the thin film transistor P21 canbe lowered. As a result, size reduction or elimination of thecomplementary capacitor C1 can be achieved. This allows the layout areato be decreased compared with the shift stages according to the otherform examples.

Moreover, in this form example, the L-level potential held by the node B(FIG. 72G) can also be lowered to VSS. Thus, the margin of the leakagetime can be widened.

The above-described features allow the shift stage according to thisform example to offer a shift stage and a shift register circuit thatare excellent in high-speed operation and operation stability and havesmall layout area.

It is also possible to use the configuration of this shift stage incombination with the configuration of any of the shift stages accordingto the above-described respective form examples.

(C) Other Form Examples (C-1) OTHER DISPLAY PANELS

The above description of the form examples is premised on application toan organic EL panel. In particular, the above description is premised onapplication to a control line driver that transfers a control pulse inthe vertical direction.

However, the above-described shift register circuit can be applied alsoto a signal line driver that gives the timings of application of thesignal potential Vsig to the signal line DTL.

Furthermore, the drive circuit including the above-described shiftregister circuit can be applied also to display panels other than theorganic EL panel.

For example, the above-described shift register circuit can be appliedalso to drive circuits in inorganic EL panels, LED panels, and otherpanels. In addition, it can be applied also to drive circuits in plasmadisplay panels. In addition, it can be applied also to drive circuits infield emission displays. Moreover, it can be applied also to drivecircuits in liquid crystal display panels. Furthermore, when the lightsource of the backlight of a liquid crystal display panel is LEDs, theshift register circuit described for the form examples can be used forthe drive circuit of the LEDs. For example, in the case of varying theratio of the lighting period in a one-field period, the shift registercircuit is suitable if the lighting period in the one-field period isdivided into plural lighting periods and the lengths and positions ofthe individual lighting periods are varied.

(C-2) PRODUCT EXAMPLES OF DISPLAY PANELS (a) Appearance Form

In this specification, the display panel encompasses not merely a panelmodule obtained by forming a pixel array part and drive circuits on aninsulating substrate by using a semiconductor process but also oneobtained by manufacturing drive circuits as circuits on a separatesubstrate (e.g. application-specific ICs) and mounting this substrate onan insulating substrate on which a pixel array part is formed.

FIG. 73 shows an appearance configuration example of a display panel. Adisplay panel 31 has a structure obtained by bonding a counter substrate35 to the formation area of a pixel array part of a support substrate33.

The support substrate 33 is composed of glass, plastic, or anotherinsulating base material (insulating substrate).

The counter substrate 35 is also composed of glass, plastic, or anotherinsulating base material (insulating substrate).

The transmittance of the substrate differs depending on the kind of thedisplay panel. For example, for a liquid crystal display panel, both thesubstrates need to have high transmittance. On the other hand, for aself-luminous display, it is sufficient that high transmittance isensured for the substrate on the light beam output side.

In the display panel 31, a flexible printed circuit (FPC) 37 for inputof external signals and drive power is disposed.

(b) Form of Mounting in Electronic Apparatus

The above-described display panel is also distributed in a form of beingmounted in various kinds of electronic apparatus. FIG. 74 shows aconceptual configuration example of electronic apparatus 41. Theelectronic apparatus 41 includes a display panel 43 including theabove-described drive circuit, a system controller 45, and an operationinput unit 47. The processing executed by the system controller 45differs depending on the commercial product form of the electronicapparatus 41. The operation input unit 47 is a device that acceptsoperation inputs to the system controller 45. As the operation inputunit 47, e.g. a mechanical interface such as a switch and a button or agraphic interface is used.

FIG. 75 shows an appearance example of a television receiver as anexample of the electronic apparatus. On the front face of the case of atelevision receiver 51, a display screen 57 composed of a front panel53, a filter glass 55, and so on is disposed. The display screen 57corresponds to the display panel 43 of FIG. 74.

Furthermore, e.g. a digital camera is available as this kind ofelectronic apparatus. FIG. 76 shows an appearance example of a digitalcamera 61. FIG. 76A shows an appearance example of the front-face side(subject side), and FIG. 76B shows an appearance example of theback-face side (photographer side).

The digital camera 61 includes a protective cover 63, an imaging lensunit 65, a display screen 67, a control switch 69, and a shutter button71. The display screen 67 corresponds to the display panel 43 of FIG.74.

Furthermore, e.g. a video camera is available as this kind of electronicapparatus. FIG. 77 shows an appearance example of a video camera 71.

The video camera 71 includes an imaging lens 75 that is disposed on thefront side of a main body 73 and used to capture an image of a subject,a start/stop switch 77 for imaging, and a display screen 79. The displayscreen 79 corresponds to the display panel 43 of FIG. 74.

Furthermore, e.g. a portable terminal device is available as this kindof electronic apparatus. FIG. 78 shows an appearance example of acellular phone 81 as the portable terminal device. The cellular phone 81shown in FIG. 78 is a foldable type. FIG. 78A shows an appearanceexample of the opened state, and FIG. 78B shows an appearance example ofthe folded state.

The cellular phone 81 includes an upper case 83, a lower case 85, aconnection (hinge, in this example) 87, a display screen 89, anauxiliary display screen 91, a picture light 93, and an imaging lens 95.The display screen 89 and the auxiliary display screen 91 correspond tothe display panel 43 of FIG. 74.

Furthermore, e.g. a computer is available as this kind of electronicapparatus. FIG. 79 shows an appearance example of a notebook computer101.

The notebook computer 101 includes a lower case 103, an upper case 105,a keyboard 107, and a display screen 109. The display screen 109corresponds to the display panel 43 of FIG. 74.

Besides the above-described devices, an audio reproduction device, agame machine, an electronic book, an electronic dictionary, and so onare available as the electronic apparatus.

(C-3) APPLICATION TO DEVICES OTHER THAN DRIVE CIRCUIT IN DISPLAY PANEL

In the above description, the shift register circuit is applied to adrive circuit that transfers a control pulse in the vertical directionof the display panel.

However, this shift register circuit can be applied also to transferringof the control pulse in the horizontal direction. Furthermore, it can beapplied to all of the shift register circuits used on a display panel.In addition, the shift register circuit is a basic circuit having highversatility and therefore can be applied to all of semiconductor devicesincluding a shift register circuit.

The present application contains subject matter related to thatdisclosed in Japanese Priority Patent Application JP 2008-096716 filedin the Japan Patent Office on Apr. 3, 2008, the entire contents of whichis hereby incorporated by reference.

(C-4) OTHERS

Various modifications might be incorporated into the above-describedform examples without departing from the scope of the present invention.In addition, various modifications and applications that are created orcombined based on the description of the present specification are alsopossible.

What is claimed is:
 1. A same-conductivity-type channel circuit,comprising: a first TFT whose current nodes are electrically connectedbetween a first input node and an output node; a second TFT whosecurrent nodes are electrically connected between a first potential lineand the output node; a third TFT whose current nodes are electricallyconnected between the first potential line and a control node of thesecond TFT; a fourth TFT whose current nodes are electrically connectedto the first potential line and the control node of the first TFT, andthe control node of the fourth TFT electrically connected to the controlnode of the second TFT; a fifth TFT whose current node is electricallyconnected to the control node of the second TFT, the control node of thefifth TFT electrically connected to a second input node; a sixth TFTwhose current node is electrically connected to the control node of thefirst TFT, the control node of the sixth TFT electrically connected to athird input node; wherein the third TFT and the fifth TFT have aconfiguration selected from a group consisting of: (i) W/L (W denoteschannel width and L denotes channel length) of the fifth TFT is equal toor larger than W/L of the third TFT; (ii) the control node of the fifthTFT is formed on only one side of a channel layer thereof, and a controlnode of the third TFT is formed on both sides of a channel layerthereof; and (iii) source-shield length of the fifth TFT is smaller thansource-shield length of the third TFT.
 2. The circuit according to claim1, wherein the W/L (W denotes channel width and L denotes channellength) of the fifth TFT is equal to or larger than the W/L of the thirdTFT.
 3. The circuit according to claim 2 wherein the W/L (W denoteschannel width and L denotes channel length) of the fifth TFT is largerthan the W/L of the third TFT.
 4. The circuit according to claim 1,wherein the control node of the fifth TFT is formed on only one side ofthe channel layer thereof, and the control node of the third TFT isformed on both sides of the channel layer thereof.
 5. The circuitaccording to claim 1, wherein the source-shield length of the fifth TFTis smaller than the source-shield length of the third TFT.
 6. Thecircuit according to claim 1, further comprising: a first capacitorelectrically connected between the control node of the first TFT and theoutput node; a second capacitor electrically connected between the firstpotential line and the control node of the second TFT.
 7. The circuitaccording to claim 1, wherein another current node of at least one ofthe fifth TFT and the sixth TFT is electrically connected to a secondpower supply line.
 8. The scanning circuit according to claim 1,wherein: the first input node is configured to receive a first timingsignal, the second input node is configured to receive a second timingsignal, and the third input node is configured to receive a third timingsignal.
 9. The circuit according to claim 1, wherein current terminalsof the fifth TFT are electrically connected between the control node ofthe second TFT and the second power supply line, and current terminalsof the sixth TFT are electrically connected between the control node ofthe first TFT and the second power supply line.
 10. The circuitaccording to claim 1, further comprising: a seventh TFT whose currentnodes are electrically connected between the first potential line andthe control node of the second TFT; and an eighth TFT whose currentnodes are electrically connected to the first potential line and thecontrol node of the first TFT.
 11. The circuit according to claim 10,wherein a control node of the seventh TFT is electrically connected tothe third input node, and a control node of the eighth TFT iselectrically connected to the second input node.
 12. The circuitaccording to claim 10, wherein the seventh TFT and the fifth TFT have aconfiguration selected from a group consisting of: (i) W/L (W denoteschannel width and L denotes channel length) of the fifth TFT is equal toor larger than W/L of the seventh TFT; (ii) the control node of thefifth TFT is formed on only one side of a channel layer thereof, and acontrol node of the seventh TFT is formed on both sides of a channellayer thereof; and (iii) source-shield length of the fifth TFT issmaller than source-shield length of the seventh TFT.
 13. A displaydevice comprising: a plurality of pixel circuits; and asame-conductivity-type channel circuit according claim 1, configured toselectively output a control signal to the pixel circuits, wherein eachof the pixel circuits includes: a switch TFT configured to receive avoltage signal; a capacitor configured to store a voltage correspondingto the voltage signal; and an electro-optical element, wherein theoutput node is coupled to control nodes of the switch transistors of thepixel circuits of a corresponding row.
 14. A display device according toclaim 13, wherein each of the pixel circuits further includes a driveTFT responsive to the capacitor; and the electro-optical elementincludes an light emitting element configured to emit light in responseto a current provided from the drive TFT.
 15. The display deviceaccording to claim 13, wherein the plurality of pixel circuits and thesame-conductivity-type channel scanning circuit are formed on a singleinsulating substrate.
 16. The display device according to claim 13,wherein the same-conductivity-type channel is a p-type channel, and theswitch TFT in each of the pixel circuits is also p-type channel TFTs.17. The display device according to claim 14, wherein thesame-conductivity-type channel is a p-type channel, and the switch TFTand a drive TFT in each of the pixel circuits are also p-type channelTFTs.
 18. The display device according to claim 13, further comprising aplurality of scanning lines, wherein the output node of thesame-conductivity-type channel circuit is coupled to control nodes ofthe switch TFTs of the pixel circuits in a corresponding row via acorresponding one of the scanning lines.